Communication system

ABSTRACT

At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into, low and high, frequency band components which are designated as a first and a second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.

This application is a Reissue of application Ser. No. 08/706,376 filedAug. 30, 1996 now U.S. Pat. No. 5,802,241 which in turn is acontinuation of application Ser. No. 08/217,895 filed Mar. 25, 1994 nowabandoned, which in turn is a Continuation-in-Part of application Ser.No. 08/126,589, filed Sep. 27, 1993, now U.S. Pat. No. 5,892,879, whichin turns is a Continuation-in-Part of application Ser. No. 08/037,108,filed Mar. 25, 1993 now U.S. Pat. No. 5,819,000. Further reissuedivisional applications have been filed, which are reissues of U.S. Pat.No. 5,802,241. These applications are: Ser. Nos. 09/698,367, filed Oct.30, 2000; 09/740,068, filed Dec. 20, 2000; and 10/301,737, filed Nov.22, 2002.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a communication system fortransmission/reception of a digital signal through modulation of itscarrier wave and demodulation of the modulated signal.

2. Description of the Prior Art

Digital signal communication systems have been used in various fields.Particularly, digital video signal transmission techniques have beenimproved remarkably.

Among them is a digital TV signal transmission method. So far, suchdigital TV signal transmission system are in particular use for e.g.transmission between TV stations. They will soon be utilized forterrestrial and/or satellite broadcast service in every country of theworld.

The TV broadcast systems including HDTV, PCM music, FAX, and otherinformation service are now demanded to increase desired data inquantity and quality for satisfying millions of sophisticated viewers.In particular, the data has to be increased in a given bandwidth offrequency allocated for TV broadcast service. The data to be transmittedis always abundant and provided as much as handled with up-to-datetechniques of the time. It is ideal to modify or change the existingsignal transmission system corresponding to an increase in the dataamount with time.

However, the TV broadcast service is a public business and cannot gofurther without considering the interests and benefits of viewers. It isessential to have any new service appreciable with existing TV receiversand displays. More particularly, the compatibility of a system is muchdesired for providing both old and new services simultaneously or onenew service which can be intercepted by either of the existing andadvanced receivers.

It is understood that any new digital TV broadcast system to beintroduced has to be arranged for data extension in order to respond tofuture demands and technological advantages and also, for compatibleaction to allow the existing receivers to receive transmissions.

The expansion capability and compatible performance of the prior artdigital TV system will be explained.

A digital satellite TV system is known in which NTSC TV signalscompressed to an about 6 Mbps are multiplexed by time divisionmodulation of 4 PSK and transmitted on 4 to 20 channels while HDTVsignals are carried on a single channel. Another digital HDTV system isprovided in which HDTV video data compressed to as small as 15 Mbps aretransmitted on a 16 or 32 QAM signal through ground stations.

Such a known satellite system permits HDTV signals to be carried on onechannel by a conventional manner, thus occupying a band of frequenciesequivalent to the same channels of NTSC signals. This causes thecorresponding NTSC channels to be unavailable during the transmission ofthe HDTV signal. Also, the compatibility between NTSC and HDTV receiversor displays is hardly concerned and data expansion capability needed formatching a future advanced mode is utterly disregarded.

Such a common terrestrial HDTV system offers and HDTV service onconventional 16 or 32 QAM signals without any modification. In anyanalogue TV broadcast service, there are developed a lot of signalattenuating or shadow regions within its service area due to structuralobstacles, geographical inconveniences, or signal interference from aneighbor station. When the TV signal is an analogue form, it can beintercepted more or less at such signal attenuating regions although itsreproduced picture is low in quality. If the TV signal is a digitalform, it can rarely be reproduced at an acceptable level within theregions. This disadvantage is critically hostile to the development ofany digital TV system.

SUMMARY OF THE INVENTION

It is an object of the present invention, for solving the foregoingdisadvantages, to provide a communication system arranged for compatibleuse for both the existing NTSC and introducing HDTV broadcast services,particularly via satellite and also, for minimizing signal attenuatingor shadow regions of its service area on the grounds ground.

The present invention provides a communication system comprising: inputmeans for inputting an image signal; image compression means forcompressing said image signal to generate a digital image compressionsignal; error correction encoding means for adding an error correctioncode to said digital image compression signal to generate an errorcorrection coding signal; modulating means for modulating said errorcorrection coding signal into an n-value VSB modulation signal;transmitting means for transmission transmitting said n-value VSBmodulation signal; receiving means for receiving a transmission signaltransmitted from said transmitting means; demodulating means fordemodulating said transmission signal into a receiving digital signal;error correction means for error correcting said receiving digitalsignal to generate an error-corrected digital signal; image expansionmeans for expanding said error-corrected digital signal to generate animage output signal; and output means for outputting said image outputsignal.

It is preferable that the n-value of the VSB modulation signal is 8.Furthermore, a Trellis decoder will be used for the error correctionmeans.

Moreover, the receiving digital signal is divided into a high prioritysignal and a low priority signal, and the error correction meansincludes a first error correction means with a high code gain and asecond error correction means with a low code gain, so that the highpriority signal is corrected by the first error correction means.

The high priority signal contains address information of data. And, theTrellis decoder will be preferable as the first error correction means.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic view of the entire arrangement of a signaltransmission system showing a first embodiment of the present invention;

FIG. 2 is a block diagram of a transmitter of the first embodiment;

FIG. 3 is a vector diagram showing a transmission signal of the firstembodiment;

FIG. 4 is a vector diagram showing a transmission signal of the firstembodiment;

FIG. 5 is a view showing an assignment of binary codes to signal pointsaccording to the first embodiment;

FIG. 6 is a view showing an assignment of binary codes to signal pointsaccording to the first embodiment;

FIG. 7 is a view showing an assignment of binary codes to signal pointsin each signal point group according to the first embodiment;

FIG. 8 is a view showing another assignment of binary codes to signalpoint groups and their signal points according to the first embodiment;

FIG. 9 is a view showing threshold values of the signal point groupsaccording to the first embodiment;

FIG. 10 is a vector diagram of a modified 16 QAM signal of the firstembodiment;

FIG. 11 is a graphic diagram showing the relation between antenna radiusr₂ and transmission energy ratio n according to the first embodiment;

FIG. 12 is a view showing the signal points of a modified 64 QAM signalof the first embodiment;

FIG. 13 is a graphic diagram showing the relation between antenna radiusr₃ and transmission energy ratio n according to the first embodiment;

FIG. 14 is a vector diagram showing signal point groups and their signalpoints of the modified 64 QAM signal of the first embodiment;

FIG. 15 is an explanatory view showing the relation between A₁ and A₂ ofthe modified 64 QAM signal of the first embodiment;

FIG. 16 is a graph diagram showing the relation between antenna radiusr₂, r₃ and transmission energy ratio n₁₆, n₆₄ respectively according tothe first embodiment;

FIG. 17 is a block diagram of a digital transmitter of the firstembodiment;

FIG. 18 is a signal space diagram of a 4 PSK modulated signal of thefirst embodiment;

FIG. 19 is a block diagram of a first receiver of the first embodiment;

FIG. 20 is a signal space diagram of a 4 PSK modulated signal of thefirst embodiment;

FIG. 21 is a block diagram of a second receiver of the first embodiment;

FIG. 22 is a vector diagram of a modified 16 QAM signal of the firstembodiment;

FIG. 23 is a vector diagram of a modified 64 QAM signal of the firstembodiment;

FIG. 24 is a flow chart showing an action of the first embodiment;

FIGS. 25(a) and 25(b) are vector diagrams showing an 8 and a 16 QAMsignal of the first embodiment respectively;

FIG. 26 is a block diagram of a third receiver of the first embodiment;

FIG. 27 is a view showing signal points of the modified 64 QAM signal ofthe first embodiment;

FIG. 28 is a flow chart showing another action of the first embodiment;

FIG. 29 is a schematic view of the entire arrangement of a signaltransmission system showing a third embodiment of the present invention;

FIG. 30 is a block diagram of a first video encoder of the thirdembodiment;

FIG. 31 is a block diagram of a first video decoder of the thirdembodiment;

FIG. 32 is a block diagram of a second video decoder of the thirdembodiment;

FIG. 33 is a block diagram of a third video decoder of the thirdembodiment;

FIG. 34 is an explanatory view showing a time multiplexing of D₁, D₂,and D₃ signals according to the third embodiment;

FIG. 35 is an explanatory view showing another time multiplexing of theD₁, D₂, and D₃ signals according to the third embodiment;

FIG. 36 is an explanatory view showing a further time multiplexing ofthe D₁, D₂, and D₃ signals according to the third embodiment;

FIG. 37 is a schematic view of the entire arrangement of a signaltransmission system showing a fourth embodiment of the presentinvention;

FIG. 38 is a vector diagram of a modified 16 QAM signal of the thirdembodiment;

FIG. 39 is a vector diagram of the modified 16 QAM signal of the thirdembodiment;

FIG. 40 is a vector diagram of a modified 64 QAM signal of the thirdembodiment;

FIG. 41 is a diagram of assignment of data components on a time baseaccording to the third embodiment;

FIG. 42 is a diagram of assignment of data components on a time base inTDMA action according to the third embodiment;

FIG. 43 is a block diagram of a carrier reproducing circuit of the thirdembodiment;

FIG. 44 is a diagram showing the principle of carrier wave reproductionaccording to the third embodiment;

FIG. 45 is a block diagram of a carrier reproducing circuit for reversemodulation of the third embodiment;

FIG. 46 is a diagram showing an assignment of signal points of the 16QAM signal of the third embodiment;

FIG. 47 is a diagram showing an assignment of signal points of the 64QAM signal of the third embodiment;

FIG. 48 is a block diagram of a carrier reproducing circuit for16×multiplication of the third embodiment;

FIG. 49 is an explanatory view showing a time multiplexing of D_(V1),D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals according to thethird embodiment;

FIG. 50 is an explanatory view showing a TDMA time multiplexing ofD_(V1), D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals according tothe third embodiment;

FIG. 51 is an explanatory view showing another TDMA time multiplexing ofthe D_(V1), D_(H1), D_(V2), D_(H2), D_(V3), and D_(H3) signals accordingto the third embodiment;

FIG. 52 is a diagram showing a signal interference region in a knowntransmission method according to the fourth embodiment;

FIG. 53 is a diagram showing signal interference regions in amulti-level signal transmission method according to the fourthembodiment;

FIG. 54 is a diagram showing signal attenuating regions in the knowntransmission method according to the fourth embodiment;

FIG. 55 is a diagram showing signal attenuating regions in themulti-level signal transmission method according to the fourthembodiment;

FIG. 56 is a diagram showing a signal interference region between twodigital TV stations according to the fourth embodiment;

FIG. 57 is a diagram showing an assignment of signal points of amodified 4 ASK signal of the fifth embodiment;

FIG. 58 is a diagram showing another assignment of signal points of themodified 4 ASK signal of the fifth embodiment;

FIGS. 59(a) to 59(d) are diagrams showing assignment of signal points ofthe modified 4 ASK signal of the fifth embodiment;

FIG. 60 is a diagram showing another assignment of signal points of themodified 4 ASK signal of the fifth embodiment when the C/N rate is low;

FIG. 61 is a block diagram of a 4 VSB or 8 VSB transmitter of the fifthembodiment;

FIG. 62(a) is a diagram showing the spectrum of an ASK signal, i.e. amulti-level VSB signal before filtering, of the fifth embodiment;

FIG. 62(b) is a diagram showing the frequency distribution profile of aVSB signal of the fifth embodiment;

FIG. 63 is a block diagram of a receiver for 4 VSB, 8 VSB and 16 VSB inaccordance with the fifth embodiment;

FIG. 64 is a block diagram of a video signal transmitter of the fifthembodiment;

FIG. 65 is a block diagram of a TV receiver of the fifth embodiment;

FIG. 66 is a block diagram of another TV receiver of the fifthembodiment;

FIG. 67 is a block diagram of a satellite-to-ground TV receiver of thefifth embodiment;

FIG. 68(a) is a diagram showing the constellation of 8 VSB of the fifthand sixth embodiments;

FIG. 68(b) is a diagram showing the constellation of 8 VSB of the fifthand sixth embodiments;

FIG. 68(c) is a view showing the signal-time waveform of 8 VSB of thefifth and sixth embodiments;

FIG. 69 is a block diagram of a video encoder of the fifth embodiment;

FIG. 70 is a block diagram of a video encoder of the fifth embodimentcontaining one divider circuit;

FIG. 71 is a block diagram of a video decoder of the fifth embodiment;

FIG. 72 is a block diagram of a video decoder of the fifth embodimentcontaining one mixer circuit;

FIG. 73 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 74(a) is a block diagram of a video decoder of the fifthembodiment;

FIG. 74(b) is a diagram showing another time assignment of datacomponents of the transmission signal according to the fifth embodiment;

FIG. 75 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 76 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 77 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 78 is a block diagram of a video decoder of the fifth embodiment;

FIG. 79 is a diagram showing a time assignment of data components of athree-level transmission signal according to the fifth embodiment;

FIG. 80 is a block diagram of another video decoder of the fifthembodiment;

FIG. 81 is a diagram showing a time assignment of data components of atransmission signal according to the fifth embodiment;

FIG. 82 is a block diagram of a video decoder for D₁ signal of the fifthembodiment;

FIG. 83 is a graphic diagram showing the relation between frequency andtime of a frequency modulated signal according to the fifth embodiment;

FIG. 84 is a block diagram of a magnetic record/playback apparatus ofthe fifth embodiment;

FIG. 85 is a graphic diagram showing the relation between C/N and levelaccording to the second embodiment;

FIG. 86 is a graphic diagram showing the relation between C/N andtransmission distance according to the second embodiment;

FIG. 87 is a block diagram of a transmission transmitter of the secondembodiment;

FIG. 88 is a block diagram of a receiver of the second embodiment;

FIG. 89 is a graphic digram showing the relation between C/N and errorrate according to the second embodiment;

FIG. 90 is a diagram showing signal attenuating regions in thethree-level transmission of the fifth embodiment;

FIG. 91 is a diagram showing signal attenuating regions in thefour-level transmission of a sixth embodiment;

FIG. 92 is a diagram showing the four-level transmission of the sixthembodiment;

FIG. 93 is a block diagram of a divider of the sixth embodiment;

FIG. 94 is a block diagram of a mixer of the sixth embodiment;

FIG. 95 is a diagram showing another four-level transmission of thesixth embodiment;

FIG. 96 is a view of signal propagation of a known digital TV broadcastsystem;

FIG. 97 is a view of signal propagation of a digital TV broadcast systemaccording to the sixth embodiment;

FIG. 98 is a diagram showing a four-level transmission of the sixthembodiment;

FIG. 99 is a vector diagram of a 16 SRQAM signal of the thirdembodiment;

FIG. 100 is a vector diagram of a 32 SRQAM signal of the thirdembodiment;

FIG. 101 is a graphic diagram showing the relation between C/N and errorrate according to the third embodiment;

FIG. 102 is a graphic diagram showing the relation between C/N and errorrate according to the third embodiment;

FIG. 103 is a graphic diagram showing the relation between shiftdistance n and C/N needed for transmission according to the thirdembodiment;

FIG. 104 is a graphic diagram showing the relation between shiftdistance n and C/N needed for transmission according to the thirdembodiment;

FIG. 105 is a graphic diagram showing the relation between signal leveland distance from a transmitter antenna in terrestrial broadcast serviceaccording to the third embodiment;

FIG. 106 is a diagram showing a service area of the 32 SRQAM signal ofthe third embodiment;

FIG. 107 is a diagram showing a service area of the 32 SRQAM signal ofthe third embodiment;

FIG. 108(a) is a diagram showing a frequency distribution profile of aconventional TV signal, FIG. 108(b) is a diagram showing a frequencydistribution profile of a conventional two-layer TV signal, FIG. 108(c)is a diagram showing threshold values of the third embodiment, FIG.108(d) is a diagram showing a frequency distribution profile oftwo-layer OFDM carriers of the ninth embodiment, and FIG. 108(e) is adiagram showing threshold values for three-layer OFDM of the ninthembodiment;

FIG. 109 is a diagram showing a time assignment of the TV signal of thethird embodiment;

FIG. 110 is a diagram showing a principle of C-CDM of the thirdembodiment;

FIG. 111 is a view showing an assignment of codes according to the thirdembodiment;

FIG. 112 is a view showing an assignment of an extended 36 QAM accordingto the third embodiment;

FIG. 113 is a view showing a frequency assignment of a modulation signalaccording to the fifth embodiment;

FIG. 114 is a block diagram showing a magnetic recording/playbackapparatus according to the fifth embodiment;

FIG. 115 is a block diagram showing a transmitter/receiver of a portabletelephone according to the eighth embodiment;

FIG. 116 is a block diagram showing base stations according to theeighth embodiment;

FIG. 117 is a view illustrating communication capacities and trafficdistribution of a conventional system;

FIG. 118 is a view illustrating communication capacities and trafficdistribution according to the eighth embodiment;

FIG. 119(a) is a diagram showing a time slot assignment of aconventional system;

FIG. 119(b) is a diagram showing a time slot assignment according to theeighth embodiment;

FIG. 120(a) is a diagram showing a time slot assignment of aconventional TDMA system;

FIG. 120(b) is a diagram showing a time slot assignment according to aTDMA system of the eighth embodiment;

FIG. 121 is a block diagram showing a one-level transmitter/receiveraccording to the eighth embodiment;

FIG. 122 is a block diagram showing a two-level transmitter/receiveraccording to the eighth embodiment;

FIG. 123 is a block diagram showing an OFDM type transmitter/receiveraccording to the ninth embodiment;

FIG. 124 is a view illustrating a principle of the OFDM system accordingto the ninth embodiment;

FIG. 125(a) is a view showing a frequency assignment of a modulationsignal of a conventional system;

FIG. 125(b) is a view showing a frequency assignment of a modulationsignal according to the ninth embodiment;

FIG. 126(a) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein no weighting is applied;

FIG. 126(b) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein two channels of two-layer OFDM areweighted by transmission electric power;

FIG. 126(c) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein carrier intervals are doubled byweighting;

FIG. 126(d) is a view showing a frequency assignment of an OFDM signalof the ninth embodiment, wherein carrier intervals are not weighted;

FIG. 127 is a block diagram showing a transmitter/receiver according tothe ninth embodiment;

FIG. 128(a) is a block diagram showing a Trellis encoder (ratio 1/2)according to the second, fourth and fifth embodiments;

FIG. 128(b) is a block diagram showing a Trellis encoder (ratio 2/3)according to the second, fourth and fifth embodiments;

FIG. 128(c) is a block diagram showing a Trellis encoder (ratio 3/4)according to the second, fourth and fifth embodiments;

FIG. 128(d) is a block diagram showing a Trellis decoder (ratio 1/2)according to the second, fourth and fifth embodiments;

FIG. 128(e) is a block diagram showing a Trellis decoder (ratio 2/3)according to the second, fourth and fifth embodiments;

FIG. 128(f) is a block diagram showing a Trellis decoder (ratio 3/4)according to the second, fourth and fifth embodiments;

FIG. 129 is a view showing a time assignment of effective symbol periodsand guard intervals according to the ninth embodiment;

FIG. 130 is a graphic diagram showing a relation between C/N rate anderror rate according to the ninth embodiment;

FIG. 131 is a block diagram showing a magnetic recording/playbackapparatus according to the fifth embodiment;

FIG. 132 is a view showing a recording format of track on the magnetictape and a travelling traveling of a head;

FIG. 133 is a block diagram showing a transmitter/receiver according tothe third embodiment;

FIG. 134 is a diagram showing a frequency assignment of a conventionalbroadcasting;

FIG. 135 is a diagram showing a relation between service area andpicture quality in a three-level signal transmission system according tothe third embodiment;

FIG. 136 is a diagram showing a frequency assignment in case themulti-level signal transmission system according to the third embodimentis combined with FDM;

FIG. 137 is a block diagram showing a transmitter/receiver according tothe third embodiment, in which Trellis encoding is adopted;

FIG. 138 is a block diagram showing a transmitter/receiver according tothe ninth embodiment, in which a part of low frequency band signal istransmitted by OFDM;

FIG. 139 is a diagram showing an assignment of signal points of the8-PS-APSK signal of the first embodiment;

FIG. 140 is a diagram showing an assignment of signal points of the16-PS-APSK signal of the first embodiment;

FIG. 141 is a diagram showing an assignment of signal points of the8-PS-PSK signal of the first embodiment;

FIG. 142 is a diagram showing an assignment of signal points of the16-PS-PSK (PS type) signal of the first embodiment;

FIG. 143 is a graphic diagram showing the relation between antennaradius of satellite and transmission capacity according to the firstembodiment;

FIG. 144 is a block diagram showing a weighted OFDM transmitter/receiveraccording to the ninth embodiment;

FIG. 145(a) is a diagram showing the waveform of the guard time and thesymbol time in the multi-level OFDM according to the ninth embodiment,wherein multipath is short;

FIG. 145(b) is a diagram showing the waveform of the guard time and thesymbol time in the multi-level OFDM according to the ninth embodiment,wherein multipath is long;

FIG. 146 is a diagram showing a principle of the multi-level OFDMaccording to the ninth embodiment;

FIG. 147 is a diagram showing subchannel assignment of a two-layersignal transmission system, weighted by electric power according to theninth embodiment;

FIG. 148 is a diagram showing relation among the D/V ratio, themultipath delay time, and the guard time according to the ninthembodiment;

FIG. 149(a) is a diagram showing time slots of respective layersaccording to the ninth embodiment;

FIG. 149(b) is a diagram showing time distribution of guard times ofrespective layers according to the ninth embodiment;

FIG. 149(c) is a diagram showing time distribution of guard times ofrespective layers according to the ninth embodiment;

FIG. 150 is a diagram showing relation between multipath delay time andtransfer rate according to the ninth embodiment, wherein three-layersignal transmission effective to multipath is realized;

FIG. 151 is a diagram showing relation between multipath delay time andC/N ratio according to the ninth embodiment, wherein two-dimensional,matrix type, multi-layer broadcast service can be realized by combiningthe GTW-OFDM and the C-CDM (or the CSW-OFDM);

FIG. 152 is a diagram showing a time assignment of a three-level TVsignal in each time slot, in the combination of the GTW-OFDM and theC-CDM (or CSW-OFDM) according to the ninth embodiment;

FIG. 153 is a diagram illustrating the multi-level broadcast systemhaving a three-dimensional matrix structure in the relationship amongthe delay time of a multipath signal, the C/N rate and the transferrate, in the combination of the GTW-OFDM and the C-CDM (or CSW-OFDM)according to the ninth embodiment;

FIG. 154 is a diagram showing the frequency distribution of thepower-weighted-OFDM according to the ninth embodiment;

FIG. 155 is a diagram showing a time assignment of a three-level TVsignal in each time slot, in the combination of the guard-time-OFDM andthe C-CDM according to the ninth embodiment;

FIG. 156 is a block diagram showing a transmitter and a receiveraccording to the fourth and fifth embodiments;

FIG. 157 is a block diagram showing a transmitter and a receiveraccording to the fourth and fifth embodiments;

FIG. 158 is a block diagram showing a transmitter and a receiveraccording to the fourth and fifth embodiments;

FIG. 159(a) is a diagram showing an assignment of signal points of the16 VSB signal of the fifth embodiment;

FIG. 159(b) is a diagram showing an assignment of signal points of the16 VSB (8 VSB) signal of the fifth embodiment;

FIG. 159(c) is a diagram showing an assignment of signal points of the16 VSB (4 VSB) signal of the fifth embodiment;

FIG. 159(d) is a diagram showing an assignment of signal points of the16 VSB (16 VSB) signal of the fifth embodiment;

FIG. 160(a) is a block circuit showing an ECC encoder according to thefifth and sixth embodiments;

FIG. 160(b) is a block circuit showing an ECC decoder according to thefifth and sixth embodiments;

FIG. 161 is a diagram showing an overall construction of a VSB receiveraccording to the fifth embodiment;

FIG. 162 is a transmitter according to the fifth embodiment;

FIG. 163 is a graph showing the relationship between the error rate andthe C/N rate with respect to the 4 VSB and the TC-8VSB;

FIG. 164 is a graph showing the relationship between the error rate andthe C/N rate with respect to the 4 VSB and the subchannels 1 and 2 ofthe TC-8VSB;

FIG. 165(a) is a block diagram showing a Reed-Solomon encoder accordingto the second, fourth and fifth embodiments;

FIG. 165(b) is a block diagram showing a Reed-Solomon decoder accordingto the second, fifth and sixth embodiments;

FIG. 166 is a flowchart showing the Reed-Solomon error correctionaccording to the second, fourth and fifth embodiment;

FIG. 167 is a block diagram showing a de interleaver according to thesecond, third, fourth, fifth and sixth embodiments;

FIG. 168(a) is a diagram showing an interleave/deinterleave tableaccording to the second, third, fourth and fifth embodiments;

FIG. 168(b) is a diagram showing an interleave distance according to thesecond, third, fourth and fifth embodiments;

FIG. 169 is a diagram showing the comparison of 4-VSB, 8-VSB and 16-VSBwith respect to the redundancy according to the fifth embodiment;

FIG. 170 is a block diagram showing a TV receiver receiving the highpriority signal according to the second, third, fourth and fifthembodiments;

FIG. 171 is a block diagram showing a transmitter and a receiveraccording to the second, third, fourth and fifth embodiments;

FIG. 172 is a block diagram showing a transmitter and a receiveraccording to the second, third, fourth and fifth embodiments; and

FIG. 173 is a block diagram showing a magnetic recording/reproducingapparatus in accordance with the ASK system of the sixth embodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Preferred embodiments of the present invention will be describedhereinafter referring to the accompanying drawings. The presentinvention can be embodied in a communication system, combining atransmitter and a receiver, for transmission/reception of a digitalsignal, such as a digital HDTV signal, and in a recording/reproducingapparatus for recording or reproducing a digital signal, such as HDTVsignal, onto or from a recording medium, such as a magnetic tape.However, construction and operating principle of the digitalmodulator/demodulator, the error-correcting encoder/decoder, and theimage (HDTV signal etc.) coding encoder/decoder of the present inventionare commonly or equally applied to each of the communication system andthe recording/reproducing apparatus. Accordingly, to describe eachembodiment efficiently, the present invention will be explained withreference to either of the communication system and therecording/reproducing apparatus. Furthermore, the present invention willbe applied to any multi-value digital modulation system which allocatessignal points on the constellation, such as QAM, ASK and PSK, althougheach embodiment may be explained based on only one modulation method.

Embodiment 1

FIG. 1 shows the entire arrangement of a signal transmission systemaccording to the first embodiment of the present invention. Atransmitter 1 comprises an input unit 2, a divider circuit 3, amodulator 4, and a transmitter unit 5. In action, each input multiplexsignal is divided by the divider circuit 3 into three groups, a firstdata stream D1, a second data stream D2, a third data stream D3, whichare then modulated by the modulator 4 before being transmitted from thetransmitter unit 5. The modulated signal is sent up from an antenna 6through an uplink 7 to a satellite 10 where it is intercepted by anuplink antenna 11 and amplified by a transponder 12 before beingtransmitted from a downlink antenna 13 towards the ground.

The transmission signal is then sent down through three downlinks 21, 3231, and 41 to a first 23, a second 33, and a third receiver 43respectively. In the first receiver 23, the signal intercepted by anantenna 22 is fed through an input unit 24 to a demodulator 25 where itsfirst data stream only is demodulated, while the second and third datastreams are not recovered, before being transmitted further from anoutput unit 26.

Similarly, the second receiver 33 allows the first and second datastreams of the signal intercepted by an antenna 32 and fed from an inputunit 34 to be demodulated by a demodulator 35 and then, summed by asummer 37 to a single data stream which is then transmitted further froman output unit 36.

The third receiver 43 allows all of the first, second, and third datastreams of the signal intercepted by an antenna 42 and fed from an inputunit 44 to be demodulated by a demodulator 45 and then, summed by asummer 47 to a single data stream which is then transmitted further froman output unit 46.

As understood, the three discrete receivers 23, 33 and 43 have theirrespective demodulators of different characteristics such that theiroutputs demodulated from the same frequency band signal of thetransmitter 1 contain data of different sizes. More particularly, threedifferent but compatible data can simultaneously be carried on a givenfrequency band signal to their respective receivers. For example, eachof three, existing NTSC, HDTV, and super HDTV, digital signals isdivided into a low, a high, and a super high frequency band componentswhich represent the first, the second, and the third data streamrespectively. Accordingly, the three different TV signals can betransmitted on a one-channel frequency band carrier for simultaneousreproduction of a medium, a high, and a super high resolution TV imagerespectively.

In service, the NTSC TV signal is intercepted by a receiver accompaniedwith a small antenna for demodulation of a small-sized data, the HDTVsignal is intercepted by a receiver accompanied with a medium antennafor demodulation of medium-sized data, and the super HDTV signal isintercepted by a receiver accompanied with a large antenna fordemodulation of large-sized data. Also, as illustrated in FIG. 1, adigital NTSC TV signal containing only the first data stream for digitalNTSC TV broadcasting service is fed to a digital transmitter 51 where itis received by an input unit 52 and modulated by a modulator 54 beforebeing transmitted further from a transmitter unit 55. The de modulatedsignal is then sent up from an antenna 56 through an uplink 57 to thesatellite 10 which in turn transmits the same through a downlink 58 tothe first receiver 23 on the ground.

The first receiver 23 demodulates with its demodulator 25 the modulateddigital signal supplied from the digital transmitter 51 to the originalfirst data stream signal. Similarly, the same modulated digital signalcan be intercepted and demodulated by the second 33 or third receiver 43to the first data stream or NTSC TV signal. In summary, the threediscrete receivers 23, 33, and 43 all can intercept and process adigital signal of the existing TV system for reproduction.

The arrangement of the signal transmission system will be described inmore detail.

FIG. 2 is a block diagram of the transmitter 1, in which an input signalis fed across the input unit 2 and divided by the divider circuit 3 intothree digital signals containing a first, a second, and a third datastream respectively.

Assuming that the input signal is a video signal, its low frequency bandcomponent is assigned to the first data stream, its high frequency bandcomponent to the second data stream, its super-high frequency bandcomponent to the third data stream. The three different frequency bandsignals are fed to a modulator input 61 of the modulator 4. Here, asignal point modulating/changing circuit 67 modulates or changes thepositions of the signal points according to an externally given signal.The modulator 4 is arranged for amplitude modulation on two90°-out-of-phase carriers respectively which are then summed to amultiple QAM signal. More specifically, the signal from the modulatorinput 61 is fed to both a first 62 and a second AM modulator 63. Also, acarrier wave of cos(2πfct) produced by a carrier generator 64 isdirectly fed to the first AM modulator 62 and also, to a π/2 phaseshifter 66 where it is 90° shifted in phase to a sin(2πfct) form priorto transmitted to the second AM modulator 63. The two amplitudemodulated signals from the first and second AM modulators 62, 63 aresummed by a summer 65 to a transmission signal which is then transferredto the transmitter unit 5 for output. The procedure is well known andwill not be further be explained.

The QAM signal will now be described in a common 8×8 4×4 or 16 stateconstellation referring to the first quadrant of a space diagram in FIG.3. The output signal of the modulator 4 is expressed by a sum vector oftwo. Acos2πfct and Bsin2πfct, vectors 81, 82 which represent the two90°-out-of-phase carriers respectively. When the distal point of a sumvector from the zero point represents a signal point, the 16 QAM signalhas 16 signal points determined by a combination of four horizontalamplitude values a₁, a₂, a₃, a₄ and four vertical amplitude values b₁,b₂, b₃, b₄. The first quadrant in FIG. 3 contains four signal points 83at C₁₁, 84 at C₁₂, 85 at C₂₂, and 86 at C₂₁.

C₁₁ is a sum vector of a vector 0-a₁ and a vector 0-b₁ and thus,expressed as C₁₁=a₁cos2πfct−b₁sin2πfct=Acos(2πfct+dπ/2).

It is now assumed that the distance between 0 and a₁, in the orthogonalcoordinates of FIG. 3 is A₁, between a₁ and a₂ is A₂, between 0 and b₁is B₁, and between b₁, and b₂ is B₂.

As shown in FIG. 4, the 16 signal points are allocated in a vectorcoordinate, in which each point represents a four-bit pattern thus toallow the transmission of four bit data per period or time slot.

FIG. 5 illustrates a common assignment of two-bit patterns to the 16signal points.

When the distance between two adjacent signal points is great, it willbe identified by the receiver with much ease. Hence, it is desired tospace the signal points at greater intervals. If two particular signalpoints are allocated near to each other, they are rarely distinguishedand the error rate will be increased. Therefore, it is most preferred tohave the signal points spaced at equal intervals as shown in FIG. 5, inwhich the 16 QAM signal is defined by A₁=A₂/2.

The transmitter 1 of the embodiment is arranged to divide an inputdigital signal into a first, a second, and a third data or bit stream.The 16 signal points or groups of signal points are divided into fourgroups. Then, 4 two-bit patterns of the first data stream are assignedto the four signal point groups respectively, as shown in FIG. 6. Moreparticularly, when the two-bit pattern of the first data stream is 11,one of four signal points of the first signal point group 91 in thefirst quadrant is selected depending on the content of the second datastream for transmission. Similarly, when 01, one signal point of thesecond signal point group 92 in the second quadrant is selected andtransmitted. When 00, one signal point of the third signal point group93 in the third quadrant is transmitted and when 10, one signal point ofthe fourth signal point group 94 in the fourth quadrant is transmitted.Also, 4 two-bit patterns in the second data stream of the 16 QAM signal,or e.g. 16 four-bit patterns in the second data stream of a 64-state QAMsignal, are assigned to four signal points or sub signal point groups ofeach of the four signal point groups 91, 92, 93, 94 respectively, asshown in FIG. 7. It should be understood that the assignment issymmetrical between any two quadrants. The assignment of the signalpoints to the four groups 91, 92, 93, 94 is determined by priority tothe two-bit data of the first data stream. As the result, two-bit dataof the first data stream and two-bit data of the second data stream canbe transmitted independently. Also, the first data stream will bedemodulated with the use of a common 4 PSK receiver having a givenantenna sensitivity. If the antenna sensitivity is higher, a modifiedtype of the 16 QAM receiver of the present invention will intercept anddemodulate both the first and second data stream streams with equalsuccess.

FIG. 8 shows an example of the assignment of the first and second datastreams in two-bit patterns.

When the low frequency band component of an HDTV video signal isassigned to the first data stream and the high frequency component tothe second data stream, the 4 PSK receiver can produce an NTSC-levelpicture from the first data stream and the 16- or 64-state QAM receivercan produce an HDTV picture from a composite reproduction signal of thefirst and second data streams.

Since the signal points are allocated at equal intervals, there isdeveloped in the 4 PSK receiver a threshold distance between thecoordinate axes and the shaded area of the first quadrant, as shown inFIG. 9. If the threshold distance is A_(T0), a PSK signal having anamplitude of A_(T0) will successfully be intercepted. However, theamplitude has to be increased to a three times greater value or 3A_(T0)for transmission of a 16 QAM signal while the threshold distance A_(T0)beingis maintained. More particularly, the energy needed fortransmitting the 16 QAM signal is nine times greater than that forsending the 4 PSK signal. Also, when the 4 PSK signal is transmitted ina 16 QAM mode, energy waste will be high and reproduction of a carriersignal will be troublesome. Above all, the energy available forsatellite transmitting is not abundant but strictly limited to minimumuse. Hence, no large-energy-consuming signal transmitting system will beput into practice until more energy for satellite transmission isavailable. It is expected that a great number of the 4 PSK receivers areintroduced into the market as digital TV broadcasting is soon inservice. After introduction to the market, the 4 PSK receivers willhardly be shifted to higher sensitivity models because a signalintercepting characteristic gap between the two, old and new, models ishigh. Therefore, the transmission of the 4 PSK signals must not beabandoned.

In this respect, a new system is desperately needed for transmitting thesignal point data of a quasi 4 PSK signal in the 16 QAM mode with theuse of less energy. Otherwise, the limited energy at a satellite stationwill degrade the entire transmission system.

The present invention resides in a multiple signal level arrangement inwhich the four signal point groups 91, 92, 93 94 are allocated at agreater distance from each other, as shown in FIG. 10, for minimizingthe energy consumption required for 16 QAM modulation of quasi 4 PSKsignals.

For clarifying the relation between the signal receiving sensitivity andthe transmitting energy, the arrangement of the digital transmitter 51and the first receiver 23 will be described in more detail referring toFIG. 1. Both the digital transmitter 51 and the first receiver 23 areformed of known types for data transmission or video signal transmissione.g. in TV broadcasting service. As shown in FIG. 17, the digitaltransmitter 51 is a 4 PSK transmitter equivalent to the multiple-bit QAMtransmitter 1, shown in FIG. 2, without AM modulation capability. Inoperation, an input signal is fed through an input unit 52 to amodulator 54 where it is divided by a modulator input 121 to twocomponents. The two components are then transferred to a first two-phasemodulator circuit 122 for phase modulation of a base carrier and asecond two-phase modulator circuit 123 for phase modulation of a carrierwhich is 90′ out of phase with the base carrier respectively. Twooutputs of the first and second two-phase modulator circuits 122, 123are then summed by a summer 65 to a composite modulated signal which isfurther transmitted from a transmitter unit 55.

The resultant modulated signal is shown in the space diagram of FIG. 18.

It is known that the four signal points are allocated at equal distancesfor achieving optimum energy utilization. FIG. 18 illustrates an examplewhere the four signal points 125, 126, 127, 128 represent 4 two-bitpatterns, 11, 01, 00, and 10 respectively. It is also desired forsuccessful data transfer from the digital transmitter 51 to the firstreceiver 23 that the 4 PSK signal from the digital transmitter 51 has anamplitude of not less than a given level. More specifically, when theminimum amplitude of the 4 PSK signal needed for transmission from thedigital transmitter 51 to the first receiver 23 of 4 PSK mode, or thedistance between 0 and a₁ in FIG. 18 is A_(T0), the first receiver 23successfully intercept intercepts any 4 PSK signal having an amplitudeof more than A_(T0).

The first receiver 23 is arranged to receive at its small-diameterantenna 22 a desired or 4 PSK signal which is transmitted from thetransmitter 1 or digital transmitter 51 respectively through thetransponder 12 of the satellite 10 and demodulate it with thedemodulator 24 25. In more particular, the first receiver 23 issubstantially designed for interception of a digital TV or datacommunications signal of 4 PSK or 2 PSK mode.

FIG. 19 is a block diagram of the first receiver 23 in which an inputsignal received by the antenna 22 from the satellite 12 10is fed throughthe input unit 24 to a carrier reproducing circuit 131 where a carrierwave is demodulated and to a π/2 phase shifter 132 where a 90° phasecarrier wave is demodulated. Also, two 90°-out-of-phase components ofthe input signal are detected by a first 133 and a second phase detectorcircuit 134 respectively and transferred to a first 136 and a seconddiscrimination/demodulation circuit 137 respectively. Two demodulatedcomponents from their respective discrimination/demodulation circuits136 and 137, which have separately been discriminated at units of timeslot by means of timing signals from a timing wave extracting circuit135, are fed to a first data stream reproducing unit 232 where they aresummed to a first data stream signal which is then delivered as anoutput from the output unit 26.

The input signal to the first receiver 23 will now be explained in moredetail referring to the vector diagram of FIG. 20. The 4 PSK signalreceived by the first receiver 23 from the digital transmitter 51 isexpressed in an ideal form without transmission distortion and noise,using four signal points 151, 152, 153, 154 shown in FIG. 20.

In practice, the real four signal points appear in particular extendedareas about the ideal signal positions 151, 152, 153, 154 respectivelydue to noise, amplitude distortion, and phase error developed duringtransmission. If one signal point is unfavorably displaced from itsoriginal position, it will hardly be distinguished from its neighborsignal point and the error rate will thus be increased. As the errorrate increases to a critical level, the reproduction of data becomesless accurate. For enabling the data reproduction at a maximumacceptable level of the error rate, the distance between any two signalpoints should be far enough to be distinguished from each other. If thedistance is 1A_(R0), the signal point 151 of a 4 PSK signal at close toa critical error level has to stay in a first discriminating area 155denoted by the hatching of FIG. 20 and determined by |0−a_(R1)|≧A_(R0)and |0−b_(R1)|≧AR0|0−b_(R1) |≧A _(R0). This allows the signaltransmission system to reproduce carrier waves and thus, demodulate awanted signal. When the minimum radius of the antenna 22 is set to r₀,the transmission signal of more than a given level can be intercepted byany receiver of the system. The amplitude of a 4 PSK signal of thedigital transmitter 51 shown in FIG. 18 is minimum at A_(T0) and thus,the minimum amplitude A_(R0) of a 4 PSK signal to be received by thefirst receiver 23 is determined to be equal to A_(T0). As the result,the first receiver 23 can intercept and demodulate the 4 PSK signal fromthe digital transmitter 51 at the maximum acceptable level of the errorrate when the radius of the antenna 22 is more than r₀. If thetransmission signal is of modified 16- or 64-state QAM mode, the firstreceiver 23 may find it difficult to reproduce its carrier wave. Forcompensation, the signal points are increased to eight which areallocated at angles of (π/4+nπ/2) as shown in FIG. 25(a) and its carrierwave will be reproduced by a 16×multiplication technique. Also, if thesignal points are assigned to 16 locations at angles of nπ/8 as shown inFIG. 25(b), the carrier of a quasi 4 PSK mode 16 QAM modulated signalcan be reproduced with the carrier reproducing circuit 131 which ismodified for performing 16×frequency multiplication. At the time, thesignal points in the transmitter 1 should be arranged to satisfyA₁/(A₁+A₂)=tan(π/8).

Here, a case of receiving a QPSK signal will be considered. SimilarlySimilar to the manner performed by the signal point modulating/changingcircuit 67 in the transmitter shown in FIG. 2, it is also possible tomodulate the positions of the signal points of the QPSK signal shown inFIG. 18 (amplitude-modulation, pulse-modulation, or the like). In thiscase, the signal point demodulating unit 138 in the first receiver 23demodulates the position modulated or position changed signal. Thedemodulated signal is outputted together with the first data stream.

The 16 PSK signal of the transmitter 1 will now be explained referringto the vector diagram of FIG. 9. When the horizontal vector distance A₁of the signal point 83 is greater than A_(T0) of the minimum amplitudeof the 4 PSK signal of the digital transmitter 51, the four signalpoints 83, 84, 85, 86 in the first quadrant of FIG. 9 stay in the shadedor first 4 PSK signal receivable area 87. When received by the firstreceiver 23, the four points of the signal appear in the firstdiscriminating area of the vector field shown in FIG. 20. Hence, any ofthe signal points 83, 84, 85, 86 of FIG. 9 can be translated into thesignal level 151 of FIG. 20 by the first receiver 23 so that the two-bitpattern of 11 is assigned to a corresponding time slot. The two-bitpattern of 11 is identical to 11 of the first signal point group 91 orfirst data stream of a signal from the transmitter 1. Equally, the firstdata stream will be reproduced at the second, third, or fourth quadrant.As the result, the first receiver 23 reproduces two-bit data of thefirst data stream out of the plurality of data streams in a 16-, 32-, or64-state QAM signal transmitted from the transmitter 1. The second andthird data streams are contained in four segments of the signal pointgroup 91 and thus, will not affect on the demodulation of the first datastream. They may however affect the reproduction of a carrier wave andan adjustment, described later, will be needed.

If the transponder of a satellite supplies an abundance of energy, theforgoing technique of 16 to 64-state QAM mode transmission will befeasible. However, the transponder of the satellite in any existingsatellite transmission system is strictly limited in the power supplydue to its compact size and the capability of solar batteries. If thetransponder or satellite is increased in size thus weight, its launchingcost will soar. This disadvantage will rarely be eliminated bytraditional techniques unless the cost of launching a satellite rocketis reduced to a considerable level. In the existing system, a commoncommunications satellite provides as low as 20 W of power supply and acommon broadcast satellite offers 100 W to 200 W at best. Fortransmission of such a 4 PSK signal in the symmetrical 16-state QAM modeas shown in FIG. 9, the minimum signal point distance is needed is3A_(T0) as the 16 QAM amplitude is expressed by 2A₁=A₂. Thus, the energyneeded for the purpose is nine times greater than that for transmissionof a common 4 PSK signal, in order to maintain compatibility. Also, anyconventional satellite transponder can hardly provide a power forenabling such a small antenna of the 4 PSK first receiver to intercept atransmitted signal therefrom. For example, in the existing 40 W system,360 W is needed for appropriate signal transmission and will beunrealistic in the respect of cost.

It would be understood that the symmetrical signal state QAM techniqueis most effective when the receivers equipped with the same sizedantennas are employed corresponding to a given transmitting power.Another novel technique will however be preferred for use with thereceivers equipped with different sized antennas.

In more detail, while the 4 PSK signal can be intercepted by a commonlow cost receiver system having a small antenna, the 16 QAM signal isintended to be received by a high cost, high quality, multiple-bitmodulating receiver system with a medium or large sized antenna which isdesigned for providing highly valuable services, e.g. HDTVentertainments, to a particular person who invests more money. Thisallows both 4 PSK and 16 QAM signals, if desired, with a 64 DMA QAM, tobe transmitted simultaneously with the help of a small increase in thetransmitting power.

For example, the transmitting power can be maintained low when thesignal points are allocated at A₁=A₂ as shown in FIG. 10. The amplitudeA(4) for transmission of 4 PSK data is expressed by a vector 96equivalent to a square root of (A₁+A₂)²+(B₁+B₂)². Then,

|A(4)) |²=A₁ ²+B₁ ²=A_(T0) ²+A_(T0) ²=2A_(T0) ²

|A(16)|²=(A₁+A₂)²+(B₁+B₂)²=4A_(T0) ²+4A_(T0) ²=8_(T0) ²

|A(16)|/|A(4)|=2

Accordingly, the 16 QAM signal can be transmitted at a two times greateramplitude and a four times greater transmitting energy than those neededfor the 4 PSK signal. A modified 16 QAM signal according to the presentinvention will not be demodulated by a common receiver designed forsymmetrical, equally distanced signal point QAM. However, it can bedemodulated with the second receiver 33 when two threshold A₁ and A₂ arepredetermined to appropriate values. At FIG. 10, the minimum distancebetween two signal points in the first segment of the signal point group91 is A₁ and A₂/2A₁ is established as compared with the distance 2A₁ of4 PSK. Then, as A₁=A₂, the distance becomes ½. This explains that thesignal receiving sensitivity has to be two times greater for the sameerror rate and four times greater for the same signal level. For havinga four times greater value of sensitivity, the radius r₂ of the antenna32 of the second receiver 33 has to be two times greater than the radiusr₁ of the antenna 22 of the first receiver 23 thus satisfying r₂=2r₁.For example, the antenna 32 of the second receiver 33 is 60 cm diameterwhen the antenna 22 of the first receiver 23 is 30 cm. In this manner,the second data stream representing the high frequency component of anHDTV will be carried on a signal channel and demodulated successfully.As the second receiver 33 intercepts the second data stream or a higherdata signal, its owner can enjoy a return of high investment. Hence, thesecond receiver 33 of a high price may be accepted. As the minimumenergy for transmission of 4 PSK data is predetermined, the ratio n₁₆ ofmodified 16 APSK transmitting energy to 4 PSK transmitting energy willbe calculated to the antenna radius r₂ of the second receiver 33 using aratio between A₁ and A₂ shown in FIG. 10.

In particular, n₁₆ is expressed by ((A₁+A₂)/A₁)² which is the minimumenergy for transmission of 4 PSK data. As the signal point distancesuited for modified 16 QAM interception is A₂, the signal point distancefor 4 PSK interception is 2A₁, and the signal point distance ratio isA₂/2A₁, the antenna radius r₂ is determined as shown in FIG. 11, inwhich the curve 101 represents the relation between the transmittingenergy ratio n₁₆ and the radius r₂ of the antenna 22 of the secondreceiver 23.

Also, the point 102 indicates transmission of common 16 QAM at the equaldistance signal state mode where the transmitting energy is nine timesgreater and thus will no more be practical. As apparent from the graphof FIG. 11, the antenna radius r₂ of the second receiver 23 cannot bereduced further even if n₁₆ is increased more than 5 times.

The transmitting energy at the satellite is limited to a small value andthus, n₁₆ preferably stays not more than 5 times the value, as denotedby the hatching of FIG. 11. The point 104 within the hatching area 103indicates, for example, that the antenna radius r₂ of a two timesgreater value is matched with a 4×value of the transmitting energy.Also, the point 105 represents that the transmission energy should bedoubled when r₂ is about 5×greater. Those values are all within afeasible range.

The value of n₁₆ not greater than 5×value is expressed using A₁ and A₂as:

n₁₆=((A₁+A₂)/A₁)²≦5

Hence, A₂≦1.23A₁.

If the distance between any two signal point group segments shown inFIG. 10 is 2A(4) and the maximum amplitude is 2A(16), A(4) andA(16)-A(4) are proportional to A₁ and A₂ respectively. Hence,(A(16))²≦5(A(14))² is established.

The action of a modified 64 ASPK transmission will be described as thethird receiver 43 can perform 64-state QAM modulation.

FIG. 12 is a vector diagram in which each signal point group segmentcontains 16 signal points as compared with 4 signal points of FIG. 10.The first signal point group segment 91 in FIG. 12 has a 4×4 matrix of16 signal points allocated at equal intervals including the point 170.For providing compatibility with 4 PSK, A₁≧A_(T0) has to be satisfied.If the radius of the antenna 42 of the third receiver 43 is r₃ and thetransmitting energy is n₆₄, the equation is expressed as:

r₃ ²={6²/(n−1)}r₁ ²

This relation between r₃ and n of a 64 QAM signal is also shown in thegraphic representation of FIG. 13.

It is understood that the signal point assignment shown in FIG. 12allows the second receiver 33 to demodulate only two-bit patterns of 4PSK data. Hence, it is desired for having compatibility between amongthe first, second, and third receivers that the second receiver 33 isarranged capable of demodulating a modified 16 QAM form from the 64 QAMmodulated signal.

The compatibility between among the three discrete receivers can beimplemented a by three-level grouping of signal points, as illustratedin FIG. 14. The description will be made referring to the first quadrantin which the first signal point group segment 91 represents the two-bitpattern 11 of the first data stream.

In particular, a first sub segment 181 in the first signal point groupsegment 91 is assigned the two-bit pattern 11 of the second data stream.Equally, a second 182, a third 183, and a fourth sub segment 184 areassigned 01, 00, and 10 of the same respectively. This assignment isidentical to that shown in FIG. 7.

The signal point allocation of the third data stream will now beexplained referring to the vector diagram of FIG. 15 which shows thefirst quadrant. As shown, the four signal points 201, 205, 209, 213represent the two-bit pattern of 11, the signal points 202, 206, 210,214 represent 01, the signal points 203, 207, 211, 215 represent 00, andsignal points 204, 208, 212, 216 represent 10. Accordingly, the two-bitpatterns of the third data stream can be transmitted separately of thefirst and second data streams. In other words, two-bit data of the threedifferent signal levels can be transmitted respectively.

As understood, the present invention permits not only transmission ofsix-bit data but also interception of three, two-bit, four-bit, andsix-bit, different bit length data with their respective receivers whilethe signal compatibility remains between among three levels.

The signal point allocation for providing compatibility between thethree levels will be described.

As shown in FIG. 15, A₁≧A_(T0) is essential for allowing the firstreceiver 23 to receive the first data stream.

It is needed to space any two signal points from each other by such adistance that the sub segment signal points, e.g. 182, 183, 184, of thesecond data stream shown in FIG. 15 can be distinguished from the signalpoint 91 shown in FIG. 10.

FIG. 15 shows that they are spaced by 2/3A₂. In this case, the distancebetween the two signal points 201 and 202 in the first sub segment 181is A₂/6. The transmitting energy needed for signal interception with thethird receiver 43 is now calculated. If the radius of the antenna 3242is r₃ and the needed transmitting energy is n₆₄ times the 4 PSKtransmitting energy, the equation is expressed as:

r₃ ²=(12r₁)²/(n−1)

This relation relationship is also denoted by the curve 221 in FIG. 16.For example, if the transmitting energy is 6 or 9 times greater thanthat for 4 PSK transmission at the point 223 or 222, the antenna 32having a radius of 8× or 6×value respectively can intercept the first,second, and third data streams for demodulation. As the signal pointdistance of the second data stream is close to 2/3A₂, the relationbetween r₁ and r₂ is expressed by:

r₂ ²=(3r₁)²/(n−1)

Therefore, the antenna 32 of the second receiver 33 has to be a littlebit increased in radius as denoted by the curve 213.

As understood, while the first and second data streams are transmittedtrough through a traditional satellite which provides a small signaltransmitting energy, the third data stream can also be transmittedthrough a future satellite which provides a greater signal transmittingenergy without interrupting the action of the first and second receivers23, 33 or with no need of modification of the same and thus, both thecompatibility and the advancement will highly be is ensured.

The signal receiving action of the second receiver 33 will first bedescribed. As compared with the first receiver 23 arranged forinterception with a small radius r₁ antenna and demodulation of the 4PSK modulated signal of the digital transmitter 51 or the first datastream of the signal of the transmitter 1, the second receiver 33 isadopted for perfectly demodulating the 16 signal state two-bit data,shown in FIG. 10, or second data stream of the 16 QAM signal from thetransmitter 1. In total, four-bit data including also the first datastream can be demodulated. The ratio between A₁ and A₂ is howeverdifferent in the two transmitters. The two different data are loaded toa demodulation controller 231 of the second receiver 33, shown in FIG.21, which in turn supplies their respective threshold values to thedemodulating circuit for AM demodulation.

The block diagram of the second receiver 33 in FIG. 21 is similar inbasic construction to that of the first receiver 23 shown in FIG. 19.The difference is that the radius r₂ of the antenna 32 is greater thanr₁ of the antenna 22. This allows the second receiver 33 to identify asignal component involving a smaller signal point distance. Thedemodulator 35 of the second receiver 33 also contains a first 232 and asecond data stream reproducing unit 233 in addition to the demodulationcontroller 231. There is provided a first discrimination/reproductioncircuit 136 for AM demodulation of modified 16 QAM signals. Asunderstood, each carrier is a four-bit signal having two, positive andnegative, threshold values about the zero level. As apparent from thevector diagram, of FIG. 22, the threshold values are varied depending onthe transmitting energy of a transmitter since the transmitting signalof the embodiment is a modified 16 QAM signal. When the referencethreshold is TH₁₆, it is determined by, as shown in FIG. 22:

TH₁₆=(A₁+A₂/2)/(A₁+A₂)

The various data for demodulation including A₁ and A₂ or TH₁₆, and thevalue m for multiple-bit modulation are also transmitted from thetransmitter 1 as carried in the first data stream. The demodulationcontroller 231 may be arranged for recovering such demodulation datathrough statistic process of the received signal.

A way of determining the shift factor A₁/A₂ will be described withreference to FIG. 26. A change of the shift factor A₁/A₂ causes a changeof the threshold value. Increase of a difference of a value of A₁/A₂ setat the receiver side from a value of A₁/A₂ set at the transmitter sidewill increase the error rate. Referring to FIG. 26, the demodulatedsignal from the second data stream reproducing unit 233 may be fed backto the demodulation controller 231 to change the shift factor A₁/A₂ in adirection to increase the error rate. By this arrangement, the thirdreceiver 43 may not demodulate the shift factor A₁/A₂, so that thecircuit construction can be simplified. Further, the transmitter may nottransmit the shift factor A₁/A₂, so that the transmission capacity canbe increased. This technique can be applied also to the second receiver33.

FIGS. 25(a) and 25(b) are views showing signal point allocations for theC-CDM signal points, wherein signal points are added by shifting in thepolar coordinate direction (r, θ). The previously described C-CDM ischaracterized in that the signal points are shifted in the rectangularcoordinate direction, i.e. XY direction; therefore it is referred to asrectangular coordinate system C-CDM. Meanwhile, this C-CDM characterizedby the shifting of signal points in the polar coordinate direction, i.e.r, θ direction, is referred to as polar coordinate system C-CDM.

FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein foursignal points are added by shifting each of 4 QPSK signals in the radiusr direction of the polar coordinate system. In this manner, the APSK ofpolar coordinate system C-CDM having 8 signal points is obtained fromthe QPSK as shown in FIG. 25(a). As the pole is shifted in the polarcoordinate system to add signal points in this APSK, it is referred toas shifted pole-APSK, i.e. SP-APSK in the abbreviated form. In thiscase, coordinate value of the newly added four QPSK signals 85 arespecified by using a shift factor S₁ as shown in FIG. 139. Namely,8PS-APSK signal points includes an include ordinary QPSK signal pointspoint 83 (r₀, θ₀) and a signal point ((S₁+1)r₀, θ₀) obtained by shiftingthe signal point 83 in the radius r direction by an amount of S₁r₀.Thus, a 1-bit subchannel 2 is obtained in addition to a 2-bit subchannel1 identical with the QPSK.

Furthermore, as shown in the constellation diagram of FIG. 140, eightnew signal points, represented by coordinates (r₀+S₂r₀, θ₀) and(r₀+S₁r₀+S₂r₀, θ₀), can be added by shifting the eight signal points(r₀, θ₀) and (r₀+S₁r₀, θ₀) in the radius r direction. As this allows twokinds of allocations, a 1-bit subchannel is obtained and is referred toas 16PS-APSK which provides the 2-bit subchannel 1, 1-bit subchannel 2,and 1-bit subchannel 3. As the 16-PS-APSK disposes the signal points onthe lines of θ=¼·(2n+1)π, it allows the ordinary QPSK receiver explainedwith reference to FIG. 19 to reproduce the carrier wave to demodulatethe first subchannel of 2-bit although the second subchannel cannot bedemodulated. As described above, the C-CDM method of shifting the signalpoints in the polar coordinate direction is useful in expanding thecapacity of information data transmission while assuring compatibilityto the PSK, especially to the QPSK receiver, a main receiver for thepresent satellite broadcast service. Therefore, without losing the firstgeneration viewers of the satellite broadcast service based on the PSK,the broadcast service will advance to a second generation stage whereinthe APSK will be used to increase transmittable information amount byuse of the multi-level modulation while maintaining compatibility.

In FIG. 25(b), the signal points are allocated on the lines of θ=π/8.With this arrangement, the 16 PSK signal points are reduced or limitedto 12 signal points, i.e. 3 signal points in each quadrant. With thislimitation, these three signal points in each quadrant are roughlyregarded as one signal point for 4 QPSK signals. Therefore this enablesthe QPSK receiver to reproduce the first subchannel in the same manneras in the previous embodiment.

More specifically, the signal points are disposed on the lines of θ=π/4,θ=π/4+π/8, and θ=π/4−π/8. In other words, the added signals are offsetby an amount ±θ in the angular direction of the polar coordinate systemfrom the QPSK signals disposed on the lines of θ=π/4. Since all thesignals are in the range of θ=π/4±π/8, they can be regarded as one ofQPSK signal points on the line of θ=π/4. Although the error rate islowered a little bit in this case, the QPSK receiver 23 shown in FIG. 19can discriminate these points as four signal points angularly allocated.Thus, 2-bit data can be reproduced.

In case of the angular shift C-CDM, if signal points are disposed on thelines of π/n, the carrier wave reproduction circuit can reproduce thecarrier wave by the use of an n-multiplier circuit in the same manner asin other embodiments. If the signal points are not disposed on the linesof π/n, the carrier wave can be reproduced by transmitting severalcarrier information within a predetermined period in the same manner asin other embodiment embodiments.

Assuming that an angle between two signal points of the QPSK or8-SP-APSK is 2θ₀ in the polar coordinate system and a first angularshift factor is P1, two signal points (r₀, θ₀+P₁θ₀) and (r₀, θ₀−P₁θ₀)are obtained by shifting the QPSK signal point in the angular θdirection by an amount ±P₁θ₀. Thus, the number of signal points isdoubled. Thus, the 1-bit subchannel 3 can be added and is referred to as8-SP-PSK of P=P1. If eight signal points are further added by shiftingthe 8-SP-PSK signals in the radius r direction by an amount S₁r₀, itwill become possible to obtain 16-SP-APSK (P, S₁ type) as shown in FIG.142. The subchannels 1 and 2 can be reproduced by two 8PS-PSKs havingthe same phase with each other. Returning to FIG. 25(b), as the C-CDMbased on the angular shift in the polar coordinate system can be appliedto the PSK as shown in FIG. 141, this will be adopted to the firstgeneration satellite broadcast service. However, if adopted to thesecond generation satellite broadcasting based on the APSK, this polarcoordinate system C-CDM is inferior in that signal points in the samegroup cannot be uniformly spaced as shown in FIG. 142. Accordingly,utilization efficiency of electric power is worsened. On the other hand,the rectangular coordinate system C-CDM has good compatibility to thePSK.

The system shown in FIG. 25(b) is compatible with both the rectangularand polar coordinate systems. As the signal points are disposed on theangular lines of the 16 PSK, they can be demodulated by the 16 PSK.Furthermore, as the signal points are divided into groups, the QPSKreceiver can be used for demodulation. Still further, as the signalpoints are also allocated to suit for the rectangular coordinate system,the demodulation will be performed by the 16-SRQAM. Consequently, thecompatibility between the rectangular coordinate system C-CDM and thepolar coordinate system C-CDM can be assured in any of the QPSK, 16PSK,and 16-SRQAM.

The demodulation controller 231 has a memory 231a for storing thereindifferent threshold values (i.e., the shift factors, the number ofsignal points, the synchronization rules, etc.) which correspond todifferent channels of TV broadcast. When receiving one of the channelsagain, the values corresponding to the receiving channel will be readout of the memory to thereby stabilize the reception quickly.

If the demodulation data is lost, the demodulation of the second datastream will hardly be executed. This will be explained referring to aflow chart shown in FIG. 24.

Even if the demodulation data is not available, demodulation of the 4PSK at Step 313 and of the first data stream at Step 301 can beimplemented. At Step 302, the demodulation data retrieved by the firstdata stream reproducing unit 232 is transferred to the demodulationcontroller 231. If m is 4 or 2 at Step 303, the demodulation controller231 triggers demodulation of 4 PSK or 2 PSK at Step 313. If not, theprocedure moves to Step 310. At Step 305, two threshold values TH₈ andTH₁₆ are calculated. The threshold value TH₁₆ for AM demodulation is fedat Step 306 from the demodulation controller 231 to both the first 136and the second discrimination/reproduction circuit 137. Hence,demodulation of the modified 16 QAM signal and reproduction of thesecond data stream can be carried out at Steps 307 and 315 respectively.At Step 308, the error rate is examined and if high, the procedurereturns to Step 313 for repeating the 4 PSK demodulation.

As shown in FIG. 22, the signal points 85, 83, are aligned on a line atan angle of cos(ωt+nπ/2) while 84 and 86 are off the line. Hence, thefeedback of a second data stream transmitting carrier wave data from thesecond data stream reproducing unit 233 to a carrier reproducing circuit131 is carried out so that no carrier needs to be extracted at thetiming of the signal points 84 and 86.

The transmitter 1 is arranged to transmit carrier timing signals atintervals of a given time with the first data stream for the purpose ofcompensation for no demodulation of the second data stream. The carriertiming signal enables to identify identification of the signal points 83and 85 of the first data stream regardless of demodulation of the seconddata stream. Hence, the reproduction of the carrier wave can betriggered by the transmitting the carrier data to the carrierreproducing circuit 131.

It is then examined at Step 304 of the flow chart of FIG. 24 whether mis 16 or not upon receipt of such a modified 64 QAM signal as shown inFIG. 23. At Step 310, it is also examined whether m is more than 64 ornot. If it is determined at Step 311 that the received signal has noequal distance signal point constellation, the procedure goes to Step312. The signal point distance TH₆₄ of the modified 64 QAM signal iscalculated from:

TH₆₄=(A₁+A₂/2)/(A₁+A₂)

This calculation is equivalent to that of TH₁₆ but its resultantdistance between signal points is smaller.

If the signal point distance in the first sub segment 181 is A₃, thedistance between the first 181 and the second sub segment 182 isexpressed by (A₂−2A₃). Then, the average distance is (A₂−2A₃)/(A₁+A₂)which is designated as d₆₄. When d₆₄ is smaller than T₂ which representsthe signal point discrimination capability of the second receiver 33,any two signal points in the segment will hardly be distinguished fromeach other. This judgement is executed at Step 313 314. If d₆₄ is out ofa permissive range, the procedure moves back to Step 313 for 4 PSK modedemodulation. If d₆₄ is within the range, the procedure advances to Step305 for allowing the demodulation of 16 QAM at Step 307. If it isdetermined at Step 308 that the error rate is too high, the proceduregoes back to Step 313 for 4 PSK mode demodulation.

When the transmitter 1 supplied a modified 8 QAM signal such as shown inFIG. 25(a) in which all the signal points are at angles ofcos(2πf+n·π/4), the carrier waves of the signal are lengthened to thesame phase and will thus be reproduced with much ease. At the time,two-bit data of the first data stream are demodulated with the 4-PSKreceiver while one-bit data of the second data stream is demodulatedwith the second receiver 33 and the total of three-bit data can bereproduced.

The third receiver 43 will be described in more detail. FIG. 26 shows ablock diagram of the third receiver 43 similar to that of the secondreceiver 33 in FIG. 21. The difference is that a third data streamreproducing unit 234 is added and also, the discrimination/reproductioncircuit has a capability of identifying eight-bit data. The antenna 42of the third receiver 43 has a radius r₃ greater than r₂ thus allowingsmaller distance state signals, e.g. 32- or 64-state QAM signals, to bedemodulated. For demodulation of the 64 QAM signal, the firstdiscrimination/reproduction circuit 136 has to identify 8 digital levelsof the detected signal in which seven different threshold levels areinvolved. As one of the threshold values is zero, three are contained inthe first quadrant.

FIG. 27 shows a space diagram of the signal in which the first quadrantcontains three different threshold values.

As shown in FIG. 27, when the three normalized threshold values areTH1₆₄, TH2₆₄, and TH3₆₄, they are expressed by:

TH1₆₄=(A₁+A₃/2)/(A₁+A₂)

TH2₆₄=(A₁+A₂/2)/(A₁+A₂)

and

TH3₆₄=(A₁+A₂−A₃/2)/(A₁+A₂)

Through AM demodulation of a phase detected signal using the threethreshold values, the third data stream can be reproduced like the firstand second data stream explained with FIG. 21. The third data streamcontains e.g. four signal points 201, 202, 203, 204 at the first subsegment 181 shown in FIG. 23 which represent 4 values of two-bitpattern. Hence, six digits or modified 64 QAM signals can bedemodulated.

The demodulation controller 231 detects the value m, A₁, A₂, and A₃ fromthe demodulation data contained in the first data stream demodulated atthe first data stream reproducing unit 232 and calculates the threethreshold values TH1₆₄, TH2₆₄, and TH3₆₄ which are then fed to the first136 and the second discrimination/reproduction circuit 137 so that themodified 64 QAM signal is demodulated with certainty. Also, if thedemodulation data have been scrambled, the modified 64 QAM signal can bedemodulated only with a specific or subscriber receiver. FIG. 28 is aflow chart showing the action of the demodulation controller 231 formodified 64 QAM signals. The difference from the flow chart fordemodulation of 16 QAM shown in FIG. 24 will be explained. The proceduremoves from Step 304 to Step 320 where it is examined whether m=32 ornot. If m=32, demodulation of 32 QAM signals is executed at Step 322. Ifnot, the procedure moves to Step 321 where it is examined whether m=64or not. If yes, A₃ is examined at Step 323. If A₃ is smaller than apredetermined value, the procedure moves to Step 305 and the samesequence as of FIG. 24 is implemented. If it is judged at Step 323 thatA₃ is not smaller than the predetermined value, the procedure goes toStep 324 where the threshold values are calculated. At Step 325, thecalculated threshold values are fed to the first and seconddiscrimination/reproduction circuits and at Step 326, the demodulationof the modified 64 QAM signal is carried out. Then, the first, second,and third data streams are reproduced at Step 327. At Step 328, theerror rate is examined. If the error rate is high, the procedure movesto Step 305 where the 16 QAM demodulation is repeated and if low, thedemodulation of the 64 QAM is continued.

The action of carrier wave reproduction needed for execution of asatisfactory demodulating procedure will now be described. The scope ofthe present invention includes reproduction of the first data stream ofa modified 16 or 64 QAM signal with the use of a 4 PSK receiver.However, a common 4 PSK receiver rarely reconstructs carrier waves, thusfailing to perform a correct demodulation. For compensation, somearrangements are necessary at both the transmitter and receiver sides.

Two techniques for the compensation are provided according to thepresent invention. A first technique relates to transmission of signalpoints aligned at angles of (2n−1)π/4 at intervals of a given time. Asecond technique offers transmission of signal points arranged atintervals of an angle of nπ/8.

According to the first technique, the eight signal points including 83and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown inFIG. 38. In action, at least one of the eight signal points istransmitted during sync time slot periods 452, 453, 454, 455 arranged atequal intervals of a time in a time slot gap 451 shown in the time chartof FIG. 38. Any desired signal points are transmitted during the othertime slots. The transmitter 1 is also arranged to assign a data for thetime slot interval to the sync timing data region 499 of a sync datablock, as shown in FIG. 41.

The content of a transmitting signal will be explained in more detailreferring to FIG. 41. The time slot group 451 containing the sync timeslots 452, 453, 454, 455 represents a unit data stream or block 491carrying a data of Dn.

The sync time slots in the signal are arranged at equal intervals of agiven time determined by the time slot interval or sync timing data.Hence, when the arrangement of the sync time slots is detected,reproduction of carrier waves will be executed slot by slot throughextracting the sync timing data from their respective time slots. Such async timing data S is contained in a sync block 493 accompanied at thefront end of a data frame 492, which consists of a number of the synctime slots denoted by the hatching in FIG. 41. Accordingly, the data tobe extracted for carrier wave reproduction are increased, thus allowingthe 4 PSK receiver to reproduce desired carrier waves at higher accuracyand efficiency.

The sync block 493 comprises sync data regions 496, 497, 498, - - -containing sync data S1, S2, S3, - - - respectively which include uniquewords and demodulation data. The phase sync signal assignment region 499is accompanied at the end of the sync block 493, which holds a data ofI_(T) including information about interval arrangement and assignment ofthe sync time slots.

The signal point data in the phase sync time slot has a particular phaseand can thus be reproduced by the 4 PSK receiver. Accordingly, I_(T) inthe phase sync signal assignment region 499 can be retrieved withouterror thus ensuring the reproduction of carrier waves at with accuracy.

As shown in FIG. 41, the sync block 493 is followed by a demodulationdata block 501 which contains demodulation data about threshold voltagesneeded for demodulation of the modified multiple-bit QAM signal. Thisdata is essential for demodulation of the multiple-bit QAM signal andmay preferably be contained in a region 502 which is a part of the syncblock 493 for ease of retrieval.

FIG. 42 shows the assignment of signal data for transmission of burstform signals through a TDMA method.

The assignment is distinguished from that of FIG. 41 by the fact that aguard period 521 is inserted between any two adjacent Dn data blocks491, 491 for interruption of the signal transmission. Also, each datablock 491 is accompanied at front end a sync region 522 thus forming adata block 492. During the sync region 522, the signal points at a phaseof (2n−1)π/4 are only transmitted. Accordingly, the carrier wavereproduction will be feasible with the 4 PSK receiver. Morespecifically, the sync signal and carrier waves can be reproducedthrough the TDMA method.

The carrier wave reproduction of the first receiver 23 shown in FIG. 19will be explained in more detail referring to FIGS. 43 and 44. As shownin FIG. 43, an input signal is fed through the input unit 24 to a syncdetector circuit 541 where it is sync detected. A demodulated signalfrom the sync detector 541 is transferred to an output circuit 542 forreproduction of the first data stream. A data of the phase sync signalassignment data region 499 (shown in FIG. 41) is retrieved with anextracting timing controller circuit 543 so that the timing of syncsignals of (2n−1)π/4 data can be acknowledged and transferred as a phasesync control pulse 561 shown in FIG. 44 to a carrier reproductioncontrolling circuit 544. Also, the demodulated signal of the syncdetector circuit 541 is fed to a frequency multiplier circuit 545 whereit is 4×multiplied prior to being transmitted to the carrierreproduction controlling circuit 544. The resultant signal denoted by562 in FIG. 44 contains a true phase data 563 and other data. Asillustrated in a time chart 564 of FIG. 44, the phase sync time slots452 carrying the (2n−1)π/4 data are also contained at equal intervals.At the carrier reproducing controlling circuit 544, the signal 562 issampled by the phase sync control pulse 561 to produce a phase samplesignal 565 which is then converted through sample-hold action to a phasesignal 566. The phase signal 566 of the carrier reproduction controllingcircuit 544 is fed across a loop filter 546 to a VCO 547 where itsrelevant carrier wave is reproduced. The reproduced carrier is then sentto the sync detector circuit 541.

In this manner, the signal point data of the (2n−1)π/4 phase denoted bythe shaded areas in FIG. 39 is recovered and utilized so that a correctcarrier wave can be reproduced by 4× or 16×frequency multiplication.Although a plurality of phases are reproduced at the time, the absolutephases of the carrier can be successfully be identified with the useduse of a unique word assigned to the sync region 496 shown in FIG. 41.

For transmission of a modified 64 QAM signal such as shown in FIG. 40,signal points in the phase sync areas 471 at the (2n−1)π/4 phase denotedby the hatching are assigned to the sync time slots 452, 452b, etc. Itscarrier can hardly be reproduced with a common 4 PSK receiver butsuccessfully with the first receiver 23 of 4 PSK mode provided with thecarrier reproducing circuit of the embodiment.

The foregoing carrier reproducing circuit is of COSTAS type. A carrierreproducing circuit of the reverse modulation type will now be explainedaccording to the embodiment.

FIG. 45 shows a reverse modulation type carrier reproducing circuitaccording to the present invention, in which a received signal is fedfrom the input unit 24 to a sync detector circuit 541 for producing ademodulated signal. Also, the input signal is delayed by a first delaycircuit 591 to a delay signal. The delay signal is then transferred to aquadrature phase modulator circuit 592 where it is reverse demodulatedby the demodulated signal from the sync detector circuit 541 to acarrier signal. The carrier signal is fed through a carrier reproductioncontroller circuit 544 to a phase comparator 593. A carrier waveproduced by a VCO 547 is delayed by a second delay circuit 594 to adelay signal which is also fed to the phase comparator 593. At the phasecomparator 594 593, the reverse demodulated carrier signal is comparedin phase with the delay signal thus producing a phase difference signal.The phase difference signal sent through a loop filter 546 to the VCO547 which in turn produces a carrier wave arranged in phase with thereceived carrier wave. In the same manner as of the COSTAS carrierreproducing circuit shown in FIG. 43, an extracting timing controllercircuit 543 performs sampling of signal points contained in the hatchingareas of FIG. 39. Accordingly, the carrier wave of a 16 or 64 QAM signalcan be reproduced with the 4 PSK demodulator of the first receiver 23.

The reproduction of a carrier wave by 16×frequency multiplication willbe explained. The transmitter 1 shown in FIG. 1 is arranged to modulateand transmit a modified 16 QAM signal with assignment of its signalpoints at nπ/8 phase as shown in FIG. 46. At the first receiver 23 shownin FIG. 19, the carrier wave can be reproduced with its COSTAS carrierreproduction controller circuit containing a 16×multiplier circuit 661shown in FIG. 48. The signal points at each nπ/8 phase shown in FIG. 46are processed at the first quadrant b by the action of the 16×multipliercircuit 661, whereby the carrier will be reproduced by the combinationof a loop filter 546 and a VCO 541 547. Also, the absolute phase may bedetermined from 16 different phases by assigning a unique word to thesync region.

The arrangement of the 16×multiplier circuit will be explained referringto FIG. 48. A sum signal and a difference signal are produced from thedemodulated signal by an adder circuit 662 and a subtractor circuit 663respectively and then, multiplied each other by a multiplier 664 to acos 2θ signal. Also, a multiplier 665 produces a sin 2θ signal. The twosignals are then multiplied by a multiplier 666 to a sin 4θ signal.

Similarly, a sin 8θ signal is produced from the two, sin 2θ and cos 2θ,signals by the combination of an adder circuit 667, a subtractor circuit668, and a multiplier multipliers 669 and 670. Furthermore, a sin 16θsignal is produced by the combination of an adder circuit 671, asubtractor circuit 672, and a multiplier 673multipliers 673 and 674.Then, the 16×multiplication is completed.

Through the foregoing 16×multiplication, the carrier wave of all thesignal points of the modified 16 QAM signal shown in FIG. 46 willsuccessfully be reproduced without extracting particular signal points.

However, reproduction of the carrier wave of the modified 64 QAM signalshown in FIG. 47 can involve an increase in the error rate due todislocation of some signal points from the sync areas 471.

Two techniques are known for compensation for the consequences. One isinhibiting transmission of the signal points dislocated from the syncareas. This causes the total amount of transmitted data to be reducedbut allows the arrangement to be facilitated. The other is providing thesync time slots as described in FIG. 38. In more particular, the signalpoints in the nπ/8 sync phase areas, e.g. 471 and 471a, are transmittedduring the period of the corresponding sync time slots in the time slotgroup 451. This triggers an accurate synchronizing action during theperiod thus minimizing phase error.

As now understood, the 16×multiplication allows the simple 4 PSKreceiver to reproduce the carrier wave of a modified 16 or 64 QAMsignal. Also, the insertion of the sync time slots causes the phasicaccuracy to be increased during the reproduction of carrier waves from amodified 64 QAM signal.

As set forth above, the signal transmission system of the presentinvention is capable of transmitting a plurality of data on a singlecarrier wave simultaneously in the multiple signal level arrangement.

More specifically, three different level receivers which have discretecharacteristics of signal intercepting sensitivity and demodulatingcapability are provided in relation to one single transmitter so thatany one of them can be selected depending on a wanted data size to bedemodulated which is proportional to the price. When the first receiverof low resolution quality and low price is acquired together with asmall antenna, its owner can intercept and reproduce the first datastream of a transmission signal. When the second receiver of mediumresolution quality and medium price is acquired together with a mediumantenna, its owner can intercept and reproduce both the first and seconddata streams of the signal. When the third receiver of high resolutionquality and high price is acquired with a large antenna, its owner canintercept and reproduce all the first, second, and third data streams ofthe signal.

If the first receiver is a home-use digital satellite broadcast receiverof low price, it will overwhelmingly be welcome by a majority ofviewers. The second receiver accompanied with the medium antenna costsmore and will be accepted by not common viewers but particular peoplewho want to enjoy HDTV services. The third receiver accompanied with thelarge antenna at least before the satellite output is increased, is notappropriate for home use and will possibly be used in relevantindustries. For example, the third data stream carrying super HDTVsignals is transmitted via a satellite to subscriber cinemas which canthus play video tapes rather than traditional movie films and run moviesbusiness at low cost.

When the present invention is applied to a TV signal transmissionservice, three different quality pictures are carried on one signalchannel wave and will offer compatibility with each other. Although thefirst embodiment refers to a 4 PSK, a modified 8 QAM, a modified 16 QAM,and a modified 64 QAM signal, other signals will also be employed withequal success including a 32 QAM, a 256 QAM, an 8 PSK, a 16 PSK, a 32PSK signal. It would be understood that the present invention is notlimited to a satellite transmission system and will be applied to aterrestrial communications system or a cable transmission system.Furthermore, the present invention will be applied to the 4- or 8-valueASK signals shown in FIGS. 58, 68(a) and 68(b).

Embodiment 2

A second embodiment of the present invention is featured in which thephysical multi-level arrangement of the first embodiment is divided intosmall levels through e.g. discrimination in error correction capability,thus forming a logic multi-level construction. In the first embodiment,each multi-level channel has different levels in the electric signalamplitude or physical demodulating capability. The second embodimentoffers different levels in the logic reproduction capability such aserror correction. For example, the data D₁ in a multi-level channel isdivided into two, D₁₋₁ and D₁₋₂, components and D₁₋₁ is more increasedin the error correction capability than D₁₋₂ for discrimination.Accordingly, as the error detection and correction capability isdifferent between D₁₋₁ and D₁₋₂ at demodulation, D₁₋₁ can successfullybe reproduced within a given error rate when the C/N level of anoriginal transmitting signal is so low as to disable the reproduction ofD₁₋₂. This will be implemented using the logic multi-level arrangement.

More specifically, the logic multi-level arrangement consists ofdividing data of a modulated multi-level channel and discriminatingdistances between error correction codes by mixing error correctioncodes with product codes for varying error correction capability. Hence,a more multi-level signal can be transmitted.

In fact, a D₁ channel is divided into two sub channels D₁₋₁ and D₁₋₂ anda D₂ channel is divided into two sub channels D₂₋₁ and D₂₋₂.

This will be explained in more detail referring to FIG. 87 85in whichD₁₋₁ is reproduced from a lowest C/N signal. If the C/N rate is d atminimum, three components D₁₋₂, D₂₋₁ and D₂₋₂ cannot be reproduced whileD₁₋₁ is reproduced. If C/N is not less than c, D₁₋₂ can also bereproduced. Equally, when C/N is b, D₂₋₁ is reproduced and when C/N isa, D₂₋₂ is reproduced. As the C/N rate increases, the reproduciblesignal levels are increased in number. The lower the C/N, the fewer thereproducible signal levels. This will be explained in the form ofrelation between transmitting distance and reproducible C/N valuereferring to FIG. 86. In common, the C/N value of a received signal isdecreased in proportion to the distance of transmission as expressed bythe real line 861 in FIG. 86. It is now assumed that the distance from atransmitter antenna to a receiver antenna is La when C/N=a, Lb whenC/N=b, Lc when C/N=c, Ld when C/N=d, and Le when C/N=e. If the distancefrom the transmitter antenna is greater than Ld, D₁₋₁ can be reproducedas shown in FIG. 85 where the receivable area 862 is denoted by thehatching. In other words, D₁₋₁ can be reproduced within a most extendedarea. Similarly, D₁₋₂ can be reproduced in an area 863 when the distanceis not more than Lc. In this area 863 containing the area 862, D₁₋₁ canwith no doubt be reproduced. In a small area 854 864, D₂₋₁ can bereproduced and in a smallest area 865, D₂₋₂ can be reproduced. Asunderstood, the different data levels of a channel can be reproducedcorresponding to degrees of declination in the C/N rate. The logicmulti-level arrangement of the signal transmission system of the presentinvention can provide the same effect as of a traditional analoguetransmission system in which the amount of receivable data is graduallylowered as the C/N rate decreases.

The construction of the logic multi-level arrangement will be describedin which there are provided two physical levels and two logic levels.FIG. 87 is a block diagram of a transmitter 1 which is substantiallyidentical in construction to that shown in FIG. 2 and describedpreviously in the first embodiment and will no further be not be furtherexplained in detail. The only difference is that error correction codeencoders are added as abbreviated to ECC encoders. The divider circuit 3has four outputs 1-1, 1-2, 2-1, and 2-2 through which four signals D₁₋₁,D₁₋₂, D₂₋₁, and D₂₋₂ divided from an input signal are delivered. The twosignals D₁₋₁ and D₁₋₂ are fed to two, main and sub, ECC encoders 872a,873a of a first ECC encoder 871a respectively for converting to errorcorrection code forms.

The main ECC encoder 872a has a higher error correction capability thanthat of the sub ECC encoder 873a. Hence, D₁₋₁ can be reproduced at alower rate of C/N than D₁₋₂ as apparent from the CN-level diagram ofFIG. 85. More particularly, the logic level of D₁₋₁ is less affected bydeclination of the C/N than that of D₁₋₂. After error correction codeencoding, D₁₋₁ and D₁₋₂ are summed by a summer 874a to a D₁ signal whichis then transferred to the modulator 4. The other two signals D₂₋₁ andD₂₋₂ of the divider circuit 3 are error correction encoded by two, mainand sub, ECC encoders 872b, 873b of a second ECC encoder 871brespectively and then, summed by a summer 874b to a D₂ signal which istransmitted to the modulator 4. The main ECC encoder 872b is higher inthe error correction capability than the sub ECC encoder 873b. Themodulator 4 in turn produces from the two, D₁ and D₂, input signals amulti-level modulated signal which is further transmitted from thetransmitter unit 5. As understood, the output signal from thetransmitter 1 has two physical levels D₁ and D₂ and also, four logiclevels D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂ based on the two physical levels forproviding different error correction capabilities.

The reception of such a multi-level signal will be explained. FIG. 88 isa block diagram of a second receiver 33 which is almost identical inconstruction to that shown in FIG. 21 and described in the firstembodiment. The second receiver 33 arranged for intercepting multi-levelsignals from the transmitter 1 shown in FIG. 87 further comprises afirst 876a and a second ECC decoder 876b, in which the demodulation ofQAM, or any of ASK, PSK, and FSK if desired, is executed.

As shown in FIG. 88, a receiver received signal is demodulated by thedemodulator 35 to the two, D₁ and D₂, signals which are then fed to twodividers 3a and 3b respectively where they are divided into four logiclevels D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂. The four signals are transferred tothe first 876a and the second ECC decoder 876b in which D₁₋₁ is errorcorrected by a main ECC decoder 877a, D₁₋₁D₁₋₂ by a sub ECC decoder878a, D₂₋₁ by a main ECC decoder 877b, D₂₋₂ by a sub ECC decoder 878bbefore all being sent to the summer 37. At the summer 37, the four,D₁₋₁, D₁₋₂, D₂₋₁, and D₂₋₂, error corrected signals are summed to asignal which is then delivered from the output unit 36.

Since D₁₋₁ and D₂₋₁ are higher in the error correction capability thanD₁₋₂ and D₂₋₂ respectively, the error rate remains less than a givenvalue although C/N is fairly low as shown in FIG. 85 and thus, anoriginal signal will be reproduced successfully.

The action of discriminating the error correction capability between themain ECC decoders 877a, 877b having high code gain and the sub ECCdecoders 878a, 878b having low code gain will now be described in moredetail. It is a good idea for having a difference in the errorcorrection capability (i.e. code gain) to use in the sub ECC decoder acommon coding technique, e.g. Reed-Solomon (FIG. 165(b)) or BCH method,having a standard code distance and in the main ECC decoder, anotherencoding technique in which the distance between correction codes isincreased using Reed-Solomon codes, their product codes, or otherlong-length codes, or Trellis decoders 744p, 744q and 744r shown inFIGS. 128(d), 128(e) and 128(f). A variety of known techniques forincreasing the error correction code distance have been introduced andwill no more explained not be explained further. The present inventioncan be associated with any known technique for having the logicmulti-level arrangement. Furthermore, as illustrated in FIGS. 160 and167, it will be preferable to provide an interleaver 744k in atransmitter and provide deinterleavers 759k, 936b in a receiver, andinterleave using an interleave table 954 of FIG. 168(a), and decode in adeinterleaver RAM 936x in the deinterleaver 936b, thereby realizing asignal transmission robust against the burst error in the transmissionsystem and therefore stabilizing images.

The logic multi-level arrangement will be explained in conjunction witha diagram of FIG. 89 showing the relation between C/N and error rateafter error correction. As shown, the straight line 881 represents D₁₋₁at the C/N and error rate relation and the line 882 represents D₁₋₂ atsame the C/N and error rate relation.

As the C/N rate of an input signal decreases, the error rate increasesafter error correction. If C/N is lower than a given value, the errorrate exceeds a reference value Eth determined by the system designstandards and no original data will normally be reconstructed. When C/Nis lowered to less than e, the D₁ signal fails to be reproduced asexpressed by the line 881 of D₁₋₁ in FIG. 89. When e≦C/N<d, D₁₋₁ of theD₁ signal exhibits a higher error rate than Eth and will not bereproduced.

When C/N is d at the point 885d, D₁₋₁ having a higher error correctioncapability than D₁₋₂ becomes not higher in the error rate than Eth andcan be reproduced. At the time, the error rate of D₁₋₂ remains higherthan Eth after error correction and will no longer be reproduced.

When C/N is increased up to c at the point 885c, D₁₋₂ becomes not higherin the error rate than Eth and can be reproduced. At the time, D₂₋₁ andD₂₋₂ remain in no demodulation state. After the C/N rate is increasedfurther to b′, the D₂ signal becomes ready to be demodulated.

When C/N is increased to b at the point 885b, D₂₋₁ of the D₂ signalbecomes not higher in the error rate than Eth and can be reproduced. Atthe time, the error rate of D₂₋₂ remains higher than Eth and will not bereproduced. When C/N is increased up to a at the point 885a, D₂₋₂becomes not higher than Eth and can be reproduced.

As described above, the four different signal logic levels divided fromtwo, D₁ and D₂, physical levels through discrimination of the errorcorrection capability between the levels, can be transmittedsimultaneously.

Using the logic multi-level arrangement of the present invention inaccompany with a multi-level construction in which at least a part ofthe original signal is reproduced even if data in a higher level islost, digital signal transmission will successfully be executed withoutlosing the advantageous effect of an analogue signal transmission inwhich transmitting data is gradually decreased as the C/N rate becomeslow.

Thanks to up-to-data date image data compression techniques, compressedimage data can be transmitted in the logic multi-level arrangement forenabling a receiver station to reproduce a higher quality image thanthat of an analogue system and also, with not sharply but at stepsdeclining the signal level for ensuring signal interception in a widerarea. The present invention can provide an extra effect of themulti-layer arrangement which is hardly implemented by a known digitalsignal transmission system without deteriorating high quality imagedata. Furthermore, the ECC encoder 743a having high code gain of FIGS.88, 133, 170 and 172 transmits, as high priority data D₁₋₁, the datamost important for image expansion of HDTV signal, such as address dataof image segment data, reference image data in the image compression,scramble release data of the descrambler of FIG. 66, and frame syncsignal. Meanwhile, the ECC decoder 758 of high code gain in the receiver43 receives this data. According to this method, even if the error rateof a signal is increased due to deterioration of the C/N ratio, theerror rate of the high priority data D₁₋₁ does no not increase so much.Therefore, it becomes possible to avoid a fatal rupture of imagepeculiar to the digital video signal, and an effect of gracefuldegradation is obtained. The modulator 749 and the demodulator 760 ofFIGS. 133 and 170 can bring the effect of graceful degradation topreviously described 16QAM, 32QAM, 4VSB of FIG. 57, 8VSB of FIG. 68, and8PSK.

Furthermore, as illustrated in FIGS. 133 and 156, the high priority datacan be error-encoded with high code gain using the ECC encoder 744a andthe Trellis encoder 744b in the 2nd data stream input 744. On the otherhand, the low priority data can be error-encoded with low code gainusing only the ECC encoder 743a. Thus, it becomes possible in the signalreception to greatly differentiate the error rate between the highpriority data and the low priority data. Therefore, the high prioritydata can be received even in the event of serious deterioration of theC/N rate in the transmission system. This is advantageous for anautomotive vehicle TV receiver or the like, whose receiving conditionsare normally unstable and the C/N ratio is bad. Although the imagequality is deteriorated with declining low priority data, the highpriority data can be surely reproduced. Therefore, allocationinformation of pixel block is reproduced and therefore the image isreconstructed without rupture, although the reproduced image isdeteriorated in the resolution and the C/N rate. Thus, the views viewerscan enjoy the TV programs.

Embodiment 3

A third embodiment of the present invention will be described referringto the relevant drawings.

FIG. 29 is a schematic total view illustrating the third embodiment inthe form of a digital TV broadcasting system. An input video signal 402of super high resolution TV image is fed to an input unit 403 of a firstvideo encoder 401. Then, the signal is divided by a divider circuit 404into three, first, second, and third, data streams which are transmittedto a compressing circuit 405 for data compression before being furtherdelivered.

Similarly, three other input video signals 406, 407, and 408 are fed toa second 409, a third 410, and a fourth video encoder 411 respectivelywhich all are arranged identical in construction to the first videoencoder 401 for data compression.

The four first data streams from their respective encoders 401, 409,410, 411 are transferred to a first multiplexer 413 of a multiplexer 412where they are time multiplexed by a TDM process to a first data streammultiplex signal which is fed to a transmitter 1.

A part or all of the four second data streams from their respectiveencoders 401, 409, 410, 411 are transferred to a second multiplexer 414of the multiplexer 412 where they are time multiplexed to a second datastream multiplex signal which is then fed to the transmitter 1. Also, apart or all of the four third data streams are transferred to a thirdmultiplexer 415 where they are time multiplexed to a third data streammultiplex signal which is then fed to the transmitter 1.

The transmitter 1 performs modulation of the three data stream signalswith its modulator 4 by the same manner as described in the firstembodiment. The modulated signals are sent from a transmitter unit 5through an antenna 6 and an uplink 7 to a transponder 12 of a satellite10 which in turn transmits it to three different receivers including afirst receiver 23.

The modulated signal transmitted through a downlink 21 is intercepted bya small antenna 22 having a radius r₁ and fed to a first data streamreproducing unit 232 of the first receiver 23 where its first datastream only is demodulated. The demodulated first data stream is thenconverted by a first video decoder 421 to a traditional 425 orwide-picture NTSC or video output signal 426 of low image resolution.

Also, the modulated signal transmitted through a downlink 31 isintercepted by a medium antenna 32 having a radius r₂ and fed to a first232 and a second data stream reproducing unit 233 of a second receiver33 where its first and second data streams are demodulated respectively.The demodulated first and second data streams are then summed andconverted by a second video decoder 422 to an HDTV or video outputsignal 427 of high image resolution and/or to the video output signals425 and 426.

Also, the modulated signal transmitted through a downlink 41 isintercepted by a large antenna 42 having a radius r₃ and fed to a first232, a second 233, and a third data stream reproducing unit 234 of athird receiver 43 where its first, second, and third data streams aredemodulated respectively. The demodulated first, second, and third datastreams are then summed and converted by a third video decoder 423 to asuper HDTV or video output signal 428 of super high image resolution foruse in a video theater or cinema. The video output signals 425, 426, and427 can also be reproduced if desired. A common signal TV signal istransmitted from a conventional digital transmitter 51 and whenintercepted by the first receiver 23, will be converted to the videooutput signal 426 such as a low resolution NTSC TV signal.

The first video encoder 401 will now be explained in more detailreferring to the block diagram of FIG. 30. An input video signal ofsuper high resolution is fed through the input unit 403 to the dividercircuit 404 where it is divided into four components by sub-band codingprocess. In more particular, the input video signal is separated throughpassing a horizontal lowpass filer 451 and a horizontal highpass filter452 of e.g. QMF QAM mode to two, low and high, horizontal frequencycomponents which are then subsampled to a half of their quantities bytwo subsamplers 453 and 454 respectively. The low horizontal componentis filtered by a vertical lowpass filter 455 and a vertical highpassfilter 456 to a low horizontal low vertical component or H_(L)V_(L)signal and a low horizontal high vertical component or H_(L)V_(H) signalrespectively. The two, H_(L)V_(L) and H_(L)V_(H), signals are thensubsampled to a half by two subsamplers 457 and 458 respectively andtransferred to the compressing circuit 405.

The high horizontal component is filtered by a vertical lowpass filter459 and a vertical highpass filter 460 to a high horizontal low verticalcomponent or H_(H)V_(L) signal and a high horizontal high verticalcomponent or H_(H)V_(H) signal respectively. The two, H_(H)V_(L) andH_(H)V_(H), signals are then subsampled to a half by two subsamplers 461and 462 respectively and transferred to the compressing circuit 405.

The H_(L)V_(L) signal is preferably DCT compressed by a first compressor471 of the compressing circuit 405 and fed to a first output 472 as thefirst data stream.

Also, the H_(L)V_(H) signal is compressed by a second compressor 473 andfed to a second output 464. The H_(H)V_(L) signal is compressed by athird compressor 463 and fed to the second output 464.

The H_(H)V_(H) signal is divided by a divider 465 into two, highresolution (H_(H)V_(H) 1) and super high resolution (H_(H)V_(H) 2),video signals which are then transferred to the second output 464 and athird output 468 respectively.

The first video decoder 421 will now be explained in more detailreferring to FIG. 31. The first data stream or D₁ signal of the firstreceiver 23 is fed through an input unit 501 to a descrambler 502 of thefirst video decoder 421 where it is descrambled. The descrambled D₁signal is expanded by an expander 503 to H_(L)V_(L) which is then fed toan aspect ratio changing circuit 504. Thus, the H_(L)V_(L) signal can bedelivered through an output unit 505 as a standard 500, letterbox format507, wide-screen 508, or sidepanel format NTSC signal 509. The scanningformat may be of non-interlace or interlace type and its NTSC mode linesmay be 525 or doubled to 1050 by double tracing. When the receivedsignal from the digital transmitter 51 is a digital TV signal of 4 PSKmode, it can also be converted by the first receiver 23 and the firstvideo decoder 421 to a TV picture. The second video decoder 422 will beexplained in more detail referring to the block diagram of FIG. 32. TheD₁ signal of the second receiver 33 is fed through a first input 521 toa first expander 522 for data expansion and then, transferred to anoversampler 523 where it is sampled at 2×. The oversampled signal isfiltered by a vertical lowpass filter 524 to H_(L)V_(L). Also, the D₂signal of the second receiver 33 is fed through a second input 530 to adivider 531 where it is divided into three components which are thentransferred to a second 532, a third 533, and a fourth expander 534respectively for data expansion. The three expanded components aresampled at 2× by three oversamplers 535, 536, 537 and filtered by avertical highpass 538, a vertical lowpass 539, and a vertical high-passfilter 540 respectively. Then, H_(L)V_(L) from the vertical lowpassfilter 524 and H_(L)V_(H) from the vertical highpass filter 538 aresummed by an adder 525, sampled by an oversampler 541, and filtered by ahorizontal lowpass filter 542 to a low frequency horizontal videosignal. H_(H)V_(L) from the vertical lowpass filter 539 and H_(H)V_(H) 1from the vertical highpass filter 540 are summed by an adder 526,sampled by an oversampler 544, and filtered by a horizontal highpassfilter 545 to a high frequency horizontal video signal. The two, highand low frequency, horizontal video signals are then summed by an adder543 to a high resolution video signal HD which is further transmittedthrough an output unit 546 as a video output 547 of e.g. HDTV format. Ifdesired a traditional NTSC video output can be reconstructed with equalsuccess.

FIG. 33 is a block diagram of the third video decoder 423 in which theD₁ and D₂ signals are fed through a first 521 and a second input 530respectively to a high frequency band video decoder circuit 527 wherethey are converted to an HD signal by the same manner as described abovedescribed . The D₃ signal is fed through a third input 551 to a superhigh frequency band video decoder circuit 552 where it is expanded,descrambled, and composed to H_(H)V_(H) 2 signal. The HD signal of thehigh frequency band video decoder circuit 527 and the H_(H)V_(H) 2signal of the super high frequency band video decoder circuit 552 aresummed by a summer 553 to a super high resolution TV or S-HD signalwhich is then delivered through an output unit 554 as a super resolutionvideo output 555.

The action of multiplexing in the multiplexer 412 shown in FIG. 29 willbe explained in more detail. FIG. 34 illustrates a data assignment inwhich the three, first, second, and third, data streams D₁, D₂, D₃contain in a period of T six NTSC channel data L1, L2, L3, L4, L5, L6,six HDTV channel data M1, M2, M3, M4, M5, M6 and six S-HDTV channel dataH1, H2, H3, H4, H5, H6 respectively. In action, the NTSC or D₁ signaldata L1 to L6 are time multiplexed by TDM process during the period T.More particularly, H_(L)V_(L) of D₁ is assigned to a domain 601 for thefirst channel. Then, a difference data M1 between HDTV and NTSC or a sumof H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) 1 is assigned to a domain 602for the first channel. Also, a difference data H1 between HDTV and superHDTV or H_(H)V_(H) 2 (See FIG. 30) is assigned to a domain 603 for thefirst channel.

The selection of the first channel TV signal will now be described. Whenintercepted by the first receiver 23 with a small antenna coupled to thefirst video decoder 421, the first channel signal is converted to astandard or widescreen NTSC TV signal as shown in FIG. 31. Whenintercepted by the second receiver 33 with a medium antenna coupled tothe second video decoder 422, the signal is converted by summing L1 ofthe first data stream D₁ assigned to the domain 601 and M1 of the seconddata stream D₂ assigned to the domain 602 to an HDTV signal of the firstchannel equivalent in program to the NTSC signal.

When intercepted by the third receiver 43 with a large antenna coupledto the third video decoder 423, the signal is converted by summing L1 ofD₁ assigned to the domain 601, M1 of D₂ assigned to the domain 602, andH₁ of D₃ assigned to the domain 603 to a super HDTV signal of the firstchannel equivalent in program to the NTSC signal. The other channelsignals can be reproduced in an equal manner.

FIG. 35 shows another data assignment L1 of a first channel NTSC signalis assigned to a first domain 601. The domain 601 which is allocated atthe front end of the first data stream D₁, also contains at front a dataS11S₁₁ including a descrambling data and the demodulation data describedin the first embodiment. A first channel HDTV signal is transmitted asL1 and M1. M1, which is thus a difference data between NTSC and HDTV, isassigned to two domains 602 and 611 of D₂. If L₁ is a compressed NTSCcomponent of 6 Mbps, M1 is as two times higher as 12 Mbps. Hence, thetotal of L1 and M1 can be demodulated at 18 Mbps with the secondreceiver 33 and the second video decoder 423. According to current datacompression techniques, HDTV compressed signals can be reproduced atabout 15 Mbps. This allows the data assignment shown in FIG. 35 toenable simultaneous reproduction of an NTSC and HDTV first channelsignal. However, this assignment allows no second channel HDTV signal tobe carried. S21 is a descrambling data in the HDTV signal. A firstchannel super HDTV signal component comprises L1, M1, and H1. Thedifference data H1 is assigned to three domains 603, 612, and 613 of D₃.If the NTSC signal is 6 Mbps, the super HDTV is carried at as high as 36Mbps. When a compressed rate is increased, super HDTV video data ofabout 2000 scanning line lines for reproduction of a cinema size picturefor commercial use can be transmitted with an equal manner.

FIG. 36 shows a further data assignment in which HI H1of a super HDTVsignal is assigned to six timestime domains. If aan NTSC compressedsignal is 6 Mbps, this assignment can carry as nine times higher as 54Mbps of D₃ data. Accordingly, super HDTV data of higher picture qualitycan be transmitted.

The foregoing data assignment makes the use of one of two, horizontaland vertical, polarization planes of a transmission wave. When both thehorizontal and vertical polarization planes are used, the frequencyutilization will be doubled. This will be explained below.

FIG. 49 shows a data assignment in which D_(V1) and D_(H1) are avertical and a horizontal polarization signal of the first data streamrespectively, D_(V2) and D_(H2) are a vertical and a horizontalpolarization signal of the second data stream respectively, and D_(V3)and D_(H3) are a vertical and a horizontal polarization signal of thethird data stream respectively. The vertical polarization signal D_(V1)of the first data stream carries a low frequency band or NTSC TV dataand the horizontal polarization signal D_(H1) carries a high frequencyband or HDTV data. When the first receiver 23 is equipped with avertical polarization antenna, it can reproduce only the NTSC signal.When the first receiver 23 is equipped with an antenna for bothhorizontally and vertically polarized waves, it can reproduce the HDTVsignal through summing L1 and M1. More specifically, the first receiver23 can provide compatibility between NTSC and HDTV with the use of aparticular type antenna.

FIG. 50 illustrates a TDMA method in which each data burst 721 isaccompanied at front by a sync data 731 and a card data 741. Also, aframe sync data 720 is provided at the front of a fame. Like channelsare assigned to like time slots. For example, a first time slot 750carries NTSC, HDTV, and super HDTV data of the first channelsimultaneously. The six time slots 750, 750a, 750b, 750c, 750d, 750e arearranged independent from each other. Hence, each station can offerNTSC, HDTV, and/or super HDTV services independently of the otherstations through selecting a particular channel of the time slots. Also,the first receiver 23 can reproduce an NTSC signal when equipped with ahorizontal polarization antenna and both NTSC and HDTV signals whenequipped with a compatible polarization antenna. In this respect, thesecond receiver 33 can reproduce a super HDTV at lower resolution whilethe third receiver 43 can reproduce a full super HDTV signal. Accordingto the third embodiment, a compatible signal transmission system will beconstructed. It is understood that the data assignment is not limited tothe burst mode TDMA method shown in FIG. 50 and another method such astime division multiplexing of continuous signals as shown in FIG. 49will be employed with equal success. Also, a data assignment shown inFIG. 51 will permit a an HDTV signal to be reproduced at highresolution.

As set forth above, the compatible digital TV signal transmission systemof the third embodiment can offer three, super HDTV, HDTV, andconventional NTSC, TV broadcast services simultaneously. In addition, avideo signal intercepted by a commercial station or cinema can beelectronized.

The modified QAM of the embodiments is now termed as SRQAM and its errorrate will be examined.

First, the error rate in 16 SRQAM will be calculated. FIG. 99 shows avector diagram of 16 SRQAM signal points. As apparent from the firstquadrant, the 16 signal points of standard 16 QAM including 83a, 83b 85,84a, 83a 86a are allocated at equal intervals of 2δ.

The signal point 83a is spaced δ from both the I-axis and the Q-axis ofthe coordinate. It is now assumed that n is a shift value of the 16SRQAM. In 16 SRQAM, the signal point 83a of 16 QAM is shifted to asignal point 83 where the distance from each axis is nδ. The shift valuen is thus expressed as:

0<n<3

The other signal points 84a and 86a are also shifted to two points 84and 86 respectively.

If the error rate of the first data stream is Pe₁, it is obtained from:$\begin{matrix}{{{Pe1} - 16} = \quad {{\frac{1}{4}\quad {erfc}\quad \left( \frac{n\quad \delta}{\sqrt{2\quad \sigma}} \right)} + {{erfc}\quad \left( \frac{3\quad \delta}{\sqrt{2\quad \sigma}} \right)}}} \\{= \quad {\frac{1}{8}\quad {erfc}\quad \left( \frac{n\sqrt{\rho}}{\sqrt{9 + n^{2}}} \right)}}\end{matrix}$

Also, the error rate Pe₂ of the second data stream is obtained from:$\begin{matrix}{{{Pe2} - 16} = \quad {\frac{1}{2}\quad {erfc}\quad \left( \frac{\frac{3 - n}{2}\quad \delta}{\sqrt{2\quad \sigma}} \right)}} \\{= \quad {\frac{1}{4}\quad {erfc}\quad \left( {\frac{3 - n}{2\sqrt{9 + n^{2}}}\quad \sqrt{\rho}} \right)}}\end{matrix}$

The error rate of 36 or 32 SRQAM will be calculated. FIG. 100 is avector diagram of a 36 SRQAM signal in which the distance between anytwo 36 QAM signal points is 28 2δ.

The signal point 83a of 36 QAM is spaced δ from each axis of thecoordinate. It is now assumed that n is a shift value of the 16 SRQAM.In 36 SRQAM, the signal point 83a is shifted to a signal point 83 wherethe distance from each axis is nδ. Similarly, the nine 36 QAM signalpoints in the first quadrant are shifted to points 83, 84, 85, 86, 97,98, 99, 100, 101 respectively. If a signal point group 90 comprising thenine signal points is regarded as a single signal point, the error ratePe₁ in reproduction of only the first data stream D₁ with a modified 4PSK receiver and the error rate Pe₂ in reproduction of the second datastream D₂ after discriminating the nine signal points of the group 90from each other, are obtained respectively from: $\begin{matrix}{{{Pe1} - 32} = \quad {\frac{1}{6}\quad {erfc}\quad \left( \frac{n\quad \delta}{\sqrt{2\quad \sigma}} \right)}} \\{= \quad {\frac{1}{6}\quad {erfc}\quad \left( {\sqrt{\frac{6p}{5}} \times \frac{n}{\sqrt{n^{2} + {2n} + 25}}} \right)}} \\{{{Pe2} - 32} = \quad {\frac{2}{3}\quad {erfc}\quad \left( \frac{5 - {n\quad \delta}}{4\sqrt{2}\quad \rho} \right)}} \\{= \quad {\frac{2}{3}\quad {erfc}\quad \left( {\sqrt{\frac{3p}{40}} \times \frac{5 - n}{\sqrt{n^{2} + {2n} + 25}}} \right)}}\end{matrix}$

FIG. 101 shows the relation between error rate Pe and C/N rate intransmission in which the curve 900 represents a conventional or notmodified 32 QAM signal. The straight line 905 represents a signal having10^(−1.5) of the error rate. The curve 901a represents a D₁ level 32SRQAM signal of the present invention at the shift rate n of 1.5. Asshown, the C/N rate of the 32 SRQAM signal is 5 dB lower at the errorrate of 10^(−1.5) than that of the conventional 32 QAM. This means thatthe present invention allows a D₁ signal to be reproduced at a givenerror rate when its C/N rate is relatively low.

The curve 902a represents a D₂ level SRQAM signal at n=1.5 which can bereproduced at the error rate of 10^(−1.5) only when its C/N rate is 2.5dB higher than that of the conventional 32 QAM of the curve 900. Also,the curves 901b and 902b represent D₁ and D₂ SRQAM signals at n=2.0respectively. The curves 902c represents a D₂ SRQAM signal at n=2.5. Itis apparent that the C/N rate of the SRQAM signal at the error rate of10^(−1.5) is 5 dB, 8 dB, and 10 dB higher at n=1.5, 2.0, and 2.5respectively in the D₁ level and 2.5 dB lower in the D₂ level than thatof a common 32 QAM signal.

Shown in FIG. 103 is the C/N rate of the first and second data streamsD₁, D₂ of a 32 SRQAM signal which is needed for maintaining a constanterror rate against variation of the shift n. As apparent, when the shiftn is more than 0.8, there is developed a clear difference between twoC/N rates of their respective D₁ and D₂ levels so that the multi-levelsignal, namely first and second data, transmission can be implementedsuccessfully. In brief, n>0.85 is essential for multi-level datatransmission of the 32 SRQAM signal of the present invention.

FIG. 102 shows the relation between the C/N rate and the error rate for16 SRQAM signals. The curve 900 represents a common 16 QAM signal. Thecurves 901a, 901b, 901c and are D₁ level or first data stream 16 SRQAMsignals at n=1.2, 1.5, and 1.8 respectively. The curves 902a, 902b, 902care D₂ level or second data stream 16 SRQAM signals at n=1.2, 1.5, and1.8 respectively.

The C/N rate of the first and second data streams D₁, D₂ of a 16 SRQAMsignal is shown in FIG. 104, which is needed for maintaining a constanterror rate against variation of the shift n. As apparent, when the shiftn is more than 0.9 (n>0.9), the multi-level data transmission of the 16SRQAM signal will be executed.

One example of propagation of SRQAM signals of the present inventionwill now be described for use with a digital TV terrestrial broadcastservice. FIG. 105 shows the relation between the signal level and thedistance between a transmitter antenna and a receiver antenna in theterrestrial broad cast service. The curve 911 represents a transmittedsignal from the transmitter antenna of 1250 feet high. It is assumedthat the error rate essential for reproduction of an applicable digitalTV signal is 10^(−1.5). The hatching area 912 represents a noiseinterruption. The point 910 represents a signal reception limit of aconventional 32 QAM signal at C/N=15 dB where the distance L is 60 milesand a digital HDTV signal can be intercepted at minimum.

The C/N rate varies 5 dB under a worse receiving condition such as badweather. If a change in the relevant condition, e.g. weather, attenuatesthe C/N rate, the interception of an HDTV signal will hardly be ensured.Also, geographical conditions largely affect the propagation of signalsand a decrease of about 10 dB at least will be unavoidable. Hence,successful signal interception within 60 miles will never be guaranteedand above all, a digital signal will be propagated harder than ananalogue signal. It would be understood that the service area of aconventional digital TV broadcast service is less dependable.

In case of the 32 SRQAM signal of the present invention, three-levelsignal transmission system is constituted as shown in FIGS. 133 and 137.This permits a low resolution NTSC signal of MPEG level to be carried onthe 1-1 data stream D₁₋₁, a medium resolution TV data of e.g. NTSCsystem to be carried on the 1-2 data stream D₁₋₂, and a high frequencycomponent of HDTV data to be carried on the second data stream D₂.Accordingly, the service area of the 1-2 data stream of the SRQAM signalis increased to a 70 mile point 910a while the service area of thesecond data stream remains within a 55 mile point 910b, as shown in FIG.105. FIG. 106 illustrates a computer simulation result of the servicearea of the 32 SRQAM signal of the present invention, which is similarto FIG. 53 but explains in more detail. As shown, the regions 708, 703c,703a, 703b, 712 represent a conventional 32 QAM receivable area, a 1-1data level D₁₋₁ receivable area, a 1-2 data level D₁₋₂ receivable area,a second data level D₂ receivable area, and a service area of a neighboranalogue TV station respectively. The conventional 32 QAM signal dataused in this drawing is based on a conventionally disclosed one.

For common 32 QAM signal, the 60-mile-radius service area can beestablished theoretically. The signal level will however be attenuatedby geographical or weather conditions and particularly, considerablydeclined at near the limit of the service area.

If the low frequency band TV component of MPEG1 grade is carried on the1-1 level D₁₋₁ data and the medium frequency band TV component of NTSCgrade on the 1-2 level D₁₋₂ data and high frequency band TV component ofHDTV on the second level D₂ data, the service area of the 32 SRQAMsignal of the present invention is increased by 10 miles in radius forreception of an EDTV signal of medium resolution grade and 18 miles forreception of an LDTV signal of low resolution grade although decreasedby 5 miles for reception of an HDTV signal of high resolution grade, asshown in FIG. 106. FIG. 107 shows a service area in case of a shiftfactor n or s=1.8. FIG. 135 shows the service area of FIG. 107 in termsof area.

More particularly, the medium resolution component of a digital TVbroadcast signal of the SRQAM mode of the preset present invention cansuccessfully be intercepted in an unfavorable service region or shadowarea where a conventional medium frequency band TV signal is hardlypropagated and attenuated due to obstacles. Within at least thepredetermined service area, the NTSC TV signal of the SRQAM mode can beintercepted by any traditional TV receiver. As the shadow or signalattenuating area developed by building structures and other obstacles orby interference of a neighbor analogue TV signal or produced in a lowland is decreased to a minimum, TV viewers or subscribers will beincreased in number.

Also, the HDTV service can be appreciated by only a few viewers who canafford to have a set of high cost HDTV receiver and display, accordingto the conventional system. The system of the present invention allows atraditional NTSC, PAL, or SECAM receiver to intercept a mediumresolution component of the digital HDTV signal with the use of anadditional digital tuner. A majority of TV viewers can hence enjoy theservice at less cost and will be increased in number. This willencourage the TV broadcast business and create an extra social benefit.

Furthermore, the signal receivable area for medium resolution or NTSC TVservice according to the present invention is increased about 36% atn=2.5, as compared with the conventional system. As the service areathus the number of TV viewers is increased, the TV broadcast businessenjoys an increasing profit. This reduces a risk in the development of anew digital TV business which will thus be encouraged to put intopractice.

FIG. 107 shows the service area of a 32 SRQAM signal of the presentinvention in which the same effect will be ensured at n=1.8. Two serviceareas 703a, 703b of D₁ and D₂ signals respectively can be determined inextension for optimum signal propagation by varying the shift nconsidering a profile of HDTV and NTSC receiver distribution orgeographical features. Accordingly, TV viewers will satisfy the serviceand a supplier station will enjoy a maximum of viewers.

This advantage is given when:

n>1.0

Hence, if the 32 SRQAM signal is selected, the shift n is determined by:

1<n<5

Also, if the 16 SRQAM signal is employed, n is determined by:

1<n<3

In the SRQAM mode signal terrestrial broadcast service in which thefirst and second data levels are created by shifting correspondingsignal points as shown in FIGS. 99 and 100, the advantage of the presentinvention will be given when the shift n in a 16, 32, or 64 SRQAM signalis more than 1.0.

In the above embodiments, the low and high frequency band components ofa video signal are transmitted as the first and second data streams.However, the transmitted signal maybe may be an audio signal. In thiscase, low frequency or low resolution components of an audio signal maybe transmitted as the first data stream, and high frequency or highresolution components of the audio signal may be transmitted as thesecond data stream. Accordingly, it is possible to receive high C/Nportion in high sound quality, and low C/N portion in low sound quality.This can be utilized in PCM broadcast, radio, portable telephone and thelike. In this case, the broadcasting area or communication distance canbe expanded as compared with the conventional systems.

Furthermore, the third embodiment can incorporate a time divisionmultiplexing (TDM) system as shown in FIG. 133. Utilization of the TDMmakes it possible to increase the number of subchannels. An ECC encoder743a and ECC encoder 743b, provided in two subchannels, differentiateECC code gains so as to make a difference between thresholds of thesetwo subchannels. Whereby, an increase of channel number of themulti-level signal transmission can be realized. In this case, it isalso possible to provide two Trellis encoders 743a, 743b as shown inFIG. 137 and differentiate their code gains. The explanation of thisblock diagram is substantially identical to that of the later describedblock diagram of FIG. 131 which shows the sixth embodiment of thepresent invention and, therefore, will not be described here.

FIG. 131 is a block diagram showing a magnetic recording/reproducingapparatus, while FIG. 137 is a block diagram showing a communicationsystem. It will be understood that these block diagrams are identicalwith each other if the up converter of the transmitter and the downconverter of the receiver in the communication system are replaced bythe magnetic head recording signal amplifier circuit and the magnetichead reproducing signal amplifier circuit in the magneticrecord/reproducing apparatus. Accordingly, construction and operation ofthe modulator and the demodulator are equivalent between these twodiagrams. In the same manner, the magnetic recording/reproducing systemof FIG. 84 is substantially identical with the communication system ofFIG. 156. The circuit of FIG. 157 will be used if simplification ofconstruction is required, and the circuit of FIG. 158 will be used ifthe simplification is further requested.

In a simulation of FIG. 106, there is provided 5 dB difference of acoding gain between 1-1 subchannel D₁₋₁ and 1-2 subchannel D₁₋₂.

An SRQAM is the system applying a C-CDM (Constellation-Code DivisionMultiplex) of the present invention to a rectangle-QAM. A C-CDM, whichis a multiplexing method independent of TDM or FDM, can obtainsubchannels by dividing a constellation-code corresponding to a code. Anincrease of the number of codes will bring an expansion of transmissioncapacity, which is not attained by TDM or FDM alone, while maintainingalmost perfect compatibility with conventional communication apparatus.Thus C-CDM can bring excellent effects.

Although above embodiment combines the C-CDM and the TDM, it is alsopossible to combine the C-CDM with the FDM (Frequency DivisionMultiplex) to obtain similar modulation effect of threshold values. Sucha system can be used for a TV broadcasting, and FIG. 108 shows afrequency distribution of a TV signal. A spectrum 725 represents afrequency distribution of a conventional analogue, e.g. NTSC,broadcasting signal. The largest signal is a video carrier 722. A colorcarrier 723 and a sound carrier 724 are not so large. There is known amethod of using an FDM for dividing a digital broadcasting signal intotwo frequencies. In this case, a carrier is divided into a first carrier726 and a second carrier 727 to transmit a first 720 and a second signal721 respectively. An interference can be lowered by placing first andsecond carriers 726, 727 sufficiently far from the video carrier 722.The first signal 720 serves to transmit a low resolution TV signal at alarge output level, while the second signal 721 serves to transmit ahigh resolution TV signal at a small output level. Consequently, themulti-level signal transmission making use of an FDM can be realizedwithout being bothered by obstruction.

FIG. 134 shows an example of a conventional method using a 32 QAMsystem. As the subchannel A has a larger output than the subchannel B, athreshold value for the subchannel A, i.e. a threshold 1, can be setsmall 4˜5 dB than a threshold value for the subchannel B, i.e. athreshold 2. Accordingly, a two-level broadcasting having 4˜5 dBthreshold difference can be realized. In this case, however, a largereduction of signal reception amount will occur if the receiving signallevel decreases below the threshold 2. Because the second signal 721a,having a large information amount as shaded in the drawing, cannot bereceived in such a case and only the first signal 720a, having a smallinformation amount, is received. Consequently, a picture quality broughtby the second level will be extremely worse.

However, the present invention resolves this problem. According to thepresent invention, the first signal 720 is given by 32 SRQAM mode whichis obtained through C-CDM modulation so that the subchannel A is dividedinto two subchannels 1 of A and 2 of A. The newly added subchannel 1 ofA, having a lowest threshold value, carries a low resolution component.The second signal 721 is also given by 32 SRQAM mode, and a thresholdvalue for the subchannel 1 of B is equalized with the threshold 2.

With this arrangement, the region in which a transmitted signal is notreceived when the signal level decreases below the threshold 2 isreduced to a shaded portion of the second signal 721a in FIG. 108. Asthe subchannel 1 of B and the subchannel A are both receivable, thetransmission amount is not so much reduced in total. Accordingly, abetter picture quality is reproduced even in the second level at thesignal level of the threshold 2.

By transmitting a normal resolution component in one subchannel, itbecomes possible to increase the number of multiple level and expand alow resolution service area. This low-threshold subchannel is utilizedfor transmitting important information such as sound information, syncinformation, headers of respective data, because these informationcarried on this low-threshold subchannel can be surely received. Thusstable reception is feasible. If a subchannel is newly added in thesecond signal 721 in the same manner, the level number of multi-leveltransmission can be increased in the service area. In the case where anHDTV signal has 1050 scanning lines, an a new service area equivalent to775 lines can be provided in addition to 525 lines.

Accordingly, the combination of the FDM and the C-CDM realizes anincrease of service area. Although the above embodiment divides asubchannel into two, it is needless to say it will also be preferable todivide it into three or more.

Next, a method of avoiding obstruction by combining the TDM and theC-CDM will be explained. As shown in FIG. 109, an analogue TV signalincludes a horizontal retrace line portion 732 and a video signalportion 731. This method utilizes a low signal level of the horizontalretrace line portion 732 and non-display of obstruction on a pictureplane during this period. By synchronizing a digital TV signal with ananalogue TV signal, horizontal retrace line sync slots 733, 733a of thehorizontal retrace line portion 732 can be used for transmission of animportant, e.g. a sync, signal or numerous data at a high output level.Thus, it becomes possible to increase data amount or output levelwithout increasing obstruction. The similar effect will be expected evenif vertical retrace line sync slots 737, 737a are provided synchronouslywith vertical retrace line portions 735, 735a.

FIG. 110 shows a principle of the C-CDM. Furthermore, FIG. 111 shows acode assignment of the C-CDM equivalent to an expanded 16 QAM. FIG. 112shows a code assignment of the C-CDM equivalent to an expanded 32 QAM.As shown in FIGS. 110 and 111, a 256 QAM signal is divided into four,740a, 740b, 740c, 740d, levels which have 4, 16, 64, 256 segments,respectively. A signal code word 742d of 256 QAM on the fourth level740d is “11111111” of 8 bit. This is split into four code words 741a,741b, 741c, and 741d of 2-bit - - - i.e. “11”, “11”, “11”, “11”, whichare then allocated on signal point regions 742a, 742b, 742c, 742d offirst, second, third, fourth levels 740a, 740b, 740c, 740d,respectively. As a result, subchannels 1, 2, 3, 4 of 2 bit are created.This is termed as C-CDM (Constellation-Code Division Multiplex). FIG.111 shows a detailed code assignment of the C-CDM equivalent to expanded16 QAM, and FIG. 112 shows a detailed code assignment of the C-CDMequivalent to expanded 32 QAM. As the C-CDM is an independentmultiplexing method, it can be combined with the conventional FDM(Frequency Division Multiplex) or TDM (Time Division Multiplex) tofurther increase the number of subchannels. In this manner, the C-CDMmethod realizes a novel multiplexing system. Although the C-CDM isexplained by using rectangle QAM, other modulation system systems havingsignal points, e.g. QAM, PSK, ASK, and even FSK if frequency regions areregarded as signal points, can be also used for this multiplexing in thesame manner.

For example, the error rate of the subchannel 1 of 8PS-APSK, explainedin the embodiment 1 with reference to FIG. 139, will be expressed asfollow follows:${{Pe1} - 8} = {{\frac{1}{4}\quad {erfc}\quad \left( \frac{\delta}{\sqrt{2}\quad \sigma} \right)} + {\frac{1}{4}\quad {erfc}\quad \left( \frac{\left( {S_{1} + 1} \right)\quad \delta}{\sqrt{2}\quad \sigma} \right)}}$

The error rate of the subchannel 2 is expressed as follows:${{Pe2} - 8} = {\frac{1}{2}\quad {erfc}\quad \left( \frac{S_{1}\quad \delta}{2\quad \sigma} \right)}$

Furthermore, the error rate of the subchannel 1 of 16-PS-APSK (PS type),explained with reference to FIG. 142, will be expressed as followfollows:${{Pe1} - 16} = {{\frac{1}{8}\quad {erfc}\quad \left( \frac{\delta}{\sqrt{2}\quad \sigma} \right)} + {\frac{1}{8}\quad {erfc}\quad \left( \frac{\left( {S_{2} + 1} \right)\quad \delta}{\sqrt{2}\quad \sigma} \right)} + {\frac{1}{8}\quad {erfc}\quad \left( \frac{\left( {S_{1} + 1} \right)\quad \delta}{\sqrt{2}\quad \sigma} \right)} + {\frac{1}{8}\quad {erfc}\quad \left( \frac{\left( {S_{1} + S_{2} + 1} \right)\quad \delta}{\sqrt{2}\quad \sigma} \right)}}$

The error rate of the subchannel 2 is expressed as follows:${{Pe2} - 16} = {{\frac{1}{4}\quad {erfc}\quad \left( \frac{S_{1}\quad \delta}{2\quad \sigma} \right)} + {\frac{1}{8}\quad {erfc}\quad \left( \frac{\left( {S_{1} - S_{2}} \right)\quad \delta}{2\quad \sigma} \right)} + {\frac{1}{8}\quad {erfc}\quad \left( \frac{\left( {S_{1} + S_{2}} \right)\quad \delta}{2\quad \sigma} \right)}}$

The error rate of the subchannel 3 is expressed as follows:${{Pe3} - 10} = {\frac{1}{2}\quad {erfc}\quad \left( \frac{S_{2}\quad \delta}{2\quad \sigma} \right)}$

Embodiment 4

A fourth embodiment of the present invention will be described referringto the relevant drawings.

FIG. 37 illustrates the entire arrangement of a signal transmissionsystem of the fourth embodiment, which is arranged for terrestrialservice and similar in both construction and action to that of the thirdembodiment shown in FIG. 29. The difference is that the transmitterantenna 6 is replaced with a terrestrial antenna 6a and the receiverantennas 22, 23, 24 32, 42are replaced with also three terrestrialantennas 22a, 23a, 24a 32a, 42a. The action of the system is identicalto that of the third embodiment and will no more be not be furtherexplained. The terrestrial broadcast service unlike a satellite servicedepends much on the distance between the transmitter antenna 6a to thereceiver antennas 22a, 32a, 42a. If a receiver is located far from thetransmitter, the level of a received signal is low. Particularly, acommon multi-level QAM signal can hardly be demodulated by the receiverwhich thus reproduces no TV program.

The signal transmission system of the present invention allows the firstreceiver 23 equipped with the antenna 22a, which is located at a fardistance as shown in FIG. 37, to intercept a modified 16 or 64 QAMsignal and demodulate at 4 PSK mode the first data stream or D₁component of the received signal to an NTSC video signal so that a TVprogram picture of medium resolution can be displayed even if the levelof the received signal is relatively low.

Also, the second receiver 33 with the antenna 32a is located at a mediumdistance from the antenna 6a and can thus intercept and demodulate boththe first and second data streams or D₁ and D₂ components of themodified 16 or 64 QAM signal to an HDTV video signal which in turnproduces an HDTV program picture.

The third receiver 43 with the antenna 42a is located at a near distanceand can intercept and demodulate the first, second, and third datastreams or D₁, D₂, and D₃ components of the modified 16 or 64 QAM signalto a super HDTV video signal which in turn produces a super HDTV picturein quality to a common movie picture.

The assignment of frequencies is determined by the same manner as of thetime division multiplexing shown in FIGS. 34, 35, and 36. Like FIG. 34,when the frequencies are assigned t to first to sixth channels, L1 ofthe D₁ component carries an NTSC data of the first channel, M1 of the D2component carries an HDTV difference data of the first channel, and H1of the D₃ component carries a super HDTV difference data of the firstchannel. Accordingly, NTSC, HDTV, and super HDTV data all can be carriedon the same channel. If D₂ and D₃ of the other channels are utilized asshown in FIGS. 35 and 36, more data of HDTV and super HDTV respectivelycan be transmitted for higher resolution display.

As understood, the system allows three different but compatible digitalTV signals to be carried on a single channel or using D₂ and D₃ regionsof other channels. Also, the medium resolution TV picture data of eachchannel can be intercepted in a wider service area according to thepresent invention.

A variety of terrestrial digital TV broadcast systems employing a 16 QAMHDTV signal of 6 MHz bandwidth have been proposed. Those are however notcompatible with the existing NTSC system and thus, have to be associatedwith a simulcast technique for transmitting NTSC signals of the sameprogram on another channel. Also, such a common 16 QAM signal limits aservice area. The terrestrial service system of the present inventionallows a receiver located at a relatively far distance to interceptsuccessfully a medium resolution TV signal with no use of an additionaldevice nor an extra channel.

FIG. 52 shows an interference region of the service area 702 of aconventional terrestrial digital HDTV broadcast station 701. As shown,the service area 702 of the conventional HDTV station 701 is intersectedwith the service area 712 of a neighbor analogue TV station 711. At theintersecting region 713, an HDTV signal is attenuated by signalinterference from the analogue TV station 711 and will thus beintercepted with less consistency.

FIG. 53 shows an interference region associated with the multi-levelsignal transmission system of the present invention. The system is lowin the energy utilization as compared with a conventional system and itsservice area 703 for HDTV signal propagation is smaller than the area702 of the conventional system. In contrary, the service area 704 fordigital NTSC or medium resolution TV signal propagation is larger thanthe conventional area 702. The level of signal interference from adigital TV station 701 of the system to a neighbor analogue TV station711 is equivalent to that from a conventional digital TV station, suchas shown in FIG. 52.

In the service area of the digital TV station 701, there are threeinterference regions developed by signal interference from the analogueTV station 711. Both HDTV and NTSC signals can hardly be intercepted inthe first region 705. Although fairly interfered, an NTSC signal may beintercepted at an equal level in the second region 706 denoted by theleft down hatching. The NTSC signal is carried on the first data streamwhich can be reproduced at a relatively low C/N rate and will thus beminimum minimally affected when the C/N rate is declined by signalinterference from the analogue TV station 711.

At the third region 707 denoted by the right down hatching, an HDTVsignal can also be intercepted when signal interference is absent whilethe NTSC signal can constantly be intercepted at a low level.

Accordingly, the overall signal receivable area of the system will beincreased although the service area of HDTV signals becomes a little bitsmaller than that of the conventional system. Also, at the signalattenuating regions produced by interference from a neighbor analogue TVstation, NTSC level signals of an HDTV program can successfully beintercepted as compared with the conventional system where no HDTVprogram is viewed in the same area. The system of the present inventionmuch reduces the size of signal attenuating area and when increases theenergy of signal transmission at a transmitter or transponder station,can extend the HDTV signal service area to an equal size to theconventional system. Also, NTSC level signals of a TV program can beintercepted more or less in a far distance area where no service isgiven by the conventional system or a signal interference area caused byan adjacent analogue TV station.

Although the embodiment employs a two-level signal transmission method,a three-level method such as shown in FIG. 78 will be used with equalsuccess. If an HDTV signal is divided into three picture levels-HDTV,NTC, and low resolution NTSC, the service area shown in FIG. 53 will beincreased from two levels to three levels where the signal propagationis extended radially and outwardly. Also, low resolution NTSC signalscan be received at an acceptable level at the first signal interferenceregion 705 where NTSC signals are hardly be intercepted in the two-levelsystem. As understood, the signal interference is also involved from adigital TV station to an analogue TV station.

The description will now be continued, provided that no digital TVstation should cause a signal interference to any neighbor analogue TVstation. According to a novel system under consideration in U.S.A.,no-use channels of the existing service channels are utilized for HDTVand thus, digital signals must not interfere with analogue signals. Forthe this purpose, the transmitting level of a digital signal has to bedecreased lower than that shown in FIG. 53. If the digital signal is ofconventional 16 QAM or 4 PSK mode, its HDTV service area 708 becomesdecreased as the signal interference region 713 denoted by the crosshatching is fairly large as shown in FIG. 54. This results in a lessnumber of viewers and sponsors, whereby such a digital system will havemuch difficulty to operate for profitable business.

FIG. 55 shows a similar result according to the system of the presentinvention. As apparent, the HDTV signal receivable area 703 is a littlebit smaller than the equal area 708 of the conventional system. However,the lower resolution or NTSC TV signal receivable area 704 will beincreased as compared with the conventional system. The hatching arearepresents a region where the NTSC level signal of a program can bereceived while the HDTV signal of the same is hardly intercepted. At thefirst interference region 705, both HDTV and NTSC signals cannot beintercepted due to signal interference from an analogue station 711.

When the level of signals is equal, the multi-level transmission systemof the present invention provides a smaller HDTV service area and agreater NTSC service area for interception of an HDTV program at an NTSCsignal level. Accordingly, the overall service area of each station isincreased and more viewers can enjoy its TV broadcasting service.Furthermore, HDTV/NTSC compatible TV business can be operated witheconomical advantages and consistency. It is also intended that thelevel of a transmitting signal is increased when the control on avertingsignal interference to neighbor analogue TV stations is lessenedcorresponding to a sharp increase in the number of home-use digitalreceivers. Hence, the service area of HDTV signals will be increased andin this respect, the two different regions for interception of HDTV/NTSCand NTSC digital TV signal levels respectively, shown in FIG. 55, can beadjusted in proportion by varying the signal point distance in the firstand/or second data stream. As the first data stream carries informationabout the signal point distance, a multi-level signal can be receivedwith more certainty.

FIG. 56 illustrates signal interference between two digital TV stationsin which a neighbor TV station 701a also provides a-digital TV broadcastservice, as compared with an analogue station in FIG. 52. Since thelevel of a transmitting signal becomes high, the HDTV service or highresolution TV signal receivable area 703inis increased to an extensionequal to the service area 702 of an analogue TV system.

At the intersecting region 714 between two service areas of theirrespective stations, the received signal can be reproduced not to anHDTV level picture with the use of a common directional antenna due tosignal interference but to an NTSC level picture with a particulardirectional antenna directed towards a desired TV station. If a highlydirectional antenna is used, the received signal from a target stationwill be reproduced to an HDTV picture. The low resolution signalreceivable area 704 is increased larger than the analogue TV systemservice area 702 and a couple of intersecting regions 715, 716 developedby the two low resolution signal receivable areas 704 and 704a of theirrespective digital TV stations 701 and 701a permit the received signalfrom antenna directed one of the two stations to be reproduced to anNTSC level picture.

The HDTV service area of the multi-level signal transmission system ofthe present invention itself will be much increased when applicablesignal restriction rules are withdrawn in a coming digital TB broadcastservice maturity time.

At the time, the system of the present invention also provides as a wideHDTV signal receivable area as of the conventional system andparticularly, allows its transmitting signal to be reproduced at an NTSClevel in a further distance or intersecting areas where TV signals ofthe conventional system are hardly intercepted. Accordingly, signalattenuating or shadow regions in the service area will be minimized.

Embodiment 5

A fifth embodiment of the present invention resides in amplitudemodulations or ASK procedure. FIG. 57 illustrates the assignment ofsignal points of a 4-level ASK signal, such as a VSB signal, accordingto the fifth embodiment, in which four signal points are denoted by 721,722, 723, and 724. FIG. 68(a) shows the constellation of the 8-value VSBsignal. The four-level transmission permits a 2-bit data to betransmitted in every cycle period, while the eight-level transmissionpermits a 4-bit data. It is assumed that the four signal points 721,722, 723, 724 represent two-bit patterns 00, 01, 10, 11, respectively,for 4-VSB.

For ease of four-level signal transmission of the embodiment, asillustrated in the signal allocation view of 4-level ASK such as 4-levelVSB of FIG. 58, the two signal points 721, 722 are designated as a firstsignal point group 725 and the other two 723, 724 are designated as asecond signal point group 726. The distance between the two signal pointgroups 725 and 726 is then determined wider than that between any twoadjacent signal points. More specifically, the distance L_(o) betweenthe two signals 722 and 723 is arranged wider than the distance Lbetween the two adjacent points 721 and 722 or 723 and 724. This isexpressed as:

L_(o)>L

Hence, the multi-level signal transmission system of the embodiment isbased on L_(o)>L. The embodiment is however not limited to L_(o)>L andL=L_(o) will be employed temporarily or permanently depending on therequirements of design, condition, and setting. FIGS. 68(a) and 68(b)show constellation for 8-value VSB.

The two signal point groups are assigned one-bit patterns of the firstdata stream D₁, as shown in FIG. 59(a). More particularly, a bit 0 ofbinary system is assigned to the first signal point group 725 andanother bit 1 to the second signal point group 726. Then, a one-bitpattern of the second data stream D₂ is assigned to each signal point.For example, the two signal points 721, 723 are assigned D₂=0 and theother two signal points 722 and 724 are assigned D₂=1. Those are thusexpressed by two bits per symbol.

The multi-level signal transmission of the present invention can beimplemented in an ASK mode with the use of the foregoing signal pointassignment. The system of the present invention works in the same manneras of a conventional equal signal point distance technique when thesignal to noise ratio or C/N rate is high. If the C/N rate becomes lowand so data can be reproduced by the conventional technique, the presentsystem ensures reproduction of the first data stream D₁ but not thesecond data stream D₂. In more detail, the state at a low C/N is shownin FIG. 60—the signal allocation diagram for ASK of 4-VSB. The signalpoints transmitted are displaced by a Gaussian distribution to ranges721a, 722a, 723a, 724a respectively at the receiver side due to noiseand transmission distortion. Therefore, the distinction between the twosignals 721 and 722 by the slice level 2 or two signals 723 and 724 bythe slice level 4 will hardly be executed. In other words, the errorrate in the second data stream D₂ will be increased. As apparent fromFIG. 60, the two signal points 721, 722 are easily distinguished fromthe other two signal points 723, 724. The distinction between the twosignal point groups 725 and 726 can thus be carried out with ease. Asthe result, the first data stream D₁ will be reproduced at a low errorrate.

Accordingly, the two different level data D₁ and D₂ can be transmittedsimultaneously. More particularly, both the first and second datastreams D₁ and D₂ of a given signal transmitted through the multi-leveltransmission system can be reproduced at the area where the C/N rate ishigh and the first data stream D₁ only can be reproduced in the areawhere the C/N rate is low.

FIG. 61 is a block diagram of a transmitter 741 in which an input unit742 comprises a first data stream input 743 and a second data streaminput 744. A carrier wave from a carrier generator 64 is amplitudemodulated by a multiplier 746 to generate 4- or 8-ASK signal as shown inFIG. 62(a) using an input signal fed across a processor 745 from theinput unit 742. The modulated 4- or 8-ASK signal is then band limited bya band-pass filter 747 to a vestigial side band having a side band witha slight residual carrier as shown in FIG. 62(b)—an ASK signal of e.g.VSB mode which is then delivered from an output unit 748.

The waveform of the ASK signal after filtering will now be examined.FIG. 62(a) shows a frequency spectrum of the ASK modulated signal inwhich two sidebands are provided on both sides of the carrier frequencyband. One of the two sidebands is eliminated with the filter 474 747toproduce a signal 749 which contains a carrier component as shown in FIG.62(b). The signal 749 is a VSB signal and if the modulation frequencyband is f_(o), will be transmitted in a frequency band of about f_(o)/2.Hence, the frequency utilization becomes high. Using VSB modetransmission, the ASK signal of two bit per symbol shown in FIG. 60 canthus carry in the same frequency band the amount of data equal to thatof 16 QAM mode at four bits per symbol for 4 VSB or that of 32 QAM modeat five bits per symbol for 8 VSB.

FIG. 63 is a block diagram of a VSB receiver 751 in which an inputsignal intercepted by a terrestrial antenna 32a is transferred throughan input unit 752 to a mixer 753 where it is mixed with a signal from avariable oscillator 754 controlled by channel selection to a lowermedium frequency signal. The signal from the mixer 753 is then detectedby a detector 755 and filtered by an LPF 756 to a baseband signal whichis transferred to a discriminating/reproduction circuit 757 having a4-level slicer for 4 VSB and an 8-level slicer for 8 VSB. Thediscrimination/reproduction circuit 757 reproduces two, first D₁ andsecond D₂, data streams from the baseband signal and transmit themfurther through a first 758 and a second data stream output 759respectively.

The transmission of a TV signal using such a transmitter and a receiverwill be explained. FIG. 64 is a block diagram of a video signaltransmitter 774 in which a high resolution TV signal, e.g. an HDTVsignal, is fed through an input unit 403 to a divider circuit 404 of afirst video encoder 401 where it is divided into four high/low frequencyTV signal components denoted by e.g. H_(L)V_(L), H_(L)V_(H), H_(H)V_(L),and H_(H)V_(H). This action is identical to that of the third embodimentpreviously described referring to FIG. 30 and will no more be not befurther explained in detail. The four separate TV signals are encodedrespectively by a compressor 405 using a known DPCMDCT variable lengthcode encoding technique which is commonly used e.g. in MPEG. Meanwhile,the motion compensation of the signal is carried out at the input unit403. The compressed signals are summed by a summer 771 to two, first andsecond, data streams D₁, D₂. The low frequency video signal component orH_(L)V_(L) signal is contained in the first data stream D₁. The two datastream signals D₁, D₂ are then transferred to a first 743 and a seconddata stream input 744 of a transmitter unit 741 where they are amplitudemodulated and summed to an ASK signal of e.g. VSB mode which ispropagated from a terrestrial antenna for broadcast service.

FIG. 65 is a block diagram of a TV receiver for such a digital TVbroadcast system. A TV broadcast signal, such as 4 VSB or 8 VSB,intercepted by a terrestrial antenna 32a is fed to an input 752 of areceiver 781. The signal is then transferred to a VSBdetection/demodulation circuit 760 where a desired channel signal isselected and demodulated to two, first and second, data streams D₁, D₂which are then fed to a first 758 768and a second data stream output 759respectively. The action in the receiver unit 751 is similar to thatdescribed previously and will no more be not be further explained indetail. The two data streams D₁, D₂ are sent to a divider unit 776 inwhich D₁ is divided by a divider 777 into two components; one orcompressed H_(L)V_(L) is transferred to a first input 521 of a secondvideo decoder 422 and the other is fed to a summer 778 where it issummed with D₂ prior to transfer to a second layer 531 of the secondvideo decoder 422. Compressed H_(L)V_(L) is then sent from the firstinput 521 to a first expander 523 where it is expanded to H_(L)V_(L) ofthe original length which is then transferred to a video mixer 548 andan aspect ratio changing circuit 779. When the input TV signal is anHDTV signal, H_(L)V_(L) represents a wide-screen NTSC signal. When thesame is an NTSC signal, H_(L)V_(L) represents a lower resolution videosignal, e.g. MPEG 1, that than an NTSC level.

The input TV signal of the embodiment is an HDTV signal and H_(L)V_(L)becomes a wide-screen NTSC signal. If the aspect ratio of an availabledisplay is 16:9, H_(L)V_(L) is directly delivered through an output unitas a 16:9 video output 426. If the display has an aspect ratio of 4:3,H_(L)V_(L) is shifted by the aspect ratio changing circuit 779 to aletterbox or sidepanel format and then, delivered from the output unit780 as a corresponding format video output 425.

The second data stream D₂ fed from the second data stream output 759 tothe summer 778 is summed with the output of the divider 777 to a sumsignal which is then fed to the second input 531 530of the second videodecoder 422. The sum signal is further transferred to a divider circuit531 while it is divided into three compressed forms of H_(L)V_(H),H_(H)V_(L), and H_(H)V_(H). The three compressed signals are then fed toa second 535, a third 536, and a fourth expander 537 respectively forconverting by expansion to H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) of theoriginal length. The three signals are summed with H_(L)V_(L) by thevideo mixer 548 to a composite HDTV signal which is fed through anoutput 546 of the second video decoder to the output unit 780. Finally,the HDTV signal is delivered from the output unit 780 as an HDTV videosignal 427.

The output unit 780 is arranged for detecting an error rate in thesecond data stream of the second data stream output 759 through an errorrate detector 782 and if the error rate is continuously high during apredetermined time, delivering H_(L)V_(L) of low resolution video datasystematically for a predetermined time.

Accordingly, the multi-level signal transmission system for digital TVsignal transmission and reception becomes feasible. For example, if a TVsignal transmitter station is near, both the first and second datastreams of a received signal can successfully be reproduced to exhibitan HDTV quality picture. If the transmitter station is far, the firstdata stream can be reproduced to H_(L)V_(L) which is converted to a lowresolution TV picture. Hence, any TV program will be intercepted in awider area and displayed at a picture quality ranging from HDTV to NTSClevel.

FIG. 66 is a block diagram showing another arrangement of the TVreceiver. As shown, the receiver unit 751 contains only a first datastream output 768 and thus, the processing of the second data stream orHDTV data is not needed so that the overall construction can beminimized. It is a good idea to have the first video decoder 421 shownin FIG. 31 as a video decoder of the receiver. Accordingly, an NTSClevel picture will be reproduced. The receiver is fabricated at muchless cost as having no capability to receive any HDTV level signal andwill widely be accepted in the market. In brief, the receiver can beused as an adapter tuner for interception of a digital TV signal withgiving no modification to the existing TV system including a display. Bythe way, in the case where the scrambled 4 VSB or 8 VSB signal isreceived as shown in FIG. 66, the scramble release signal transmitted bythe 4 VSB or 8 VSB signal is collated with the number of the descramblenumber memory 502c in the descrambler 502 by the descramble numberchecker 502b. Only when the scramble release signal is identical withthe number of the descramble number memory 502c, the scramble of aspecific scramble program can be duly released by releasing descramble.

The TV receiver 781 may have a further arrangement shown in FIG. 67,which serves as both a satellite broadcast receiver for demodulation ofPSK signals and a terrestrial broadcast receiver for demodulation of VSBsignals. In action, a PSK signal received by a satellite antenna 32 ismixed by a mixer 786 with a signal from an oscillator 787 to a lowfrequency signal which is then fed through an input unit 34 to a mixer753 similar to one shown in FIG. 63. The low frequency signal of PSK orQAM mode in a given channel of the satellite TV system is transferred toa modulator demodulator 35 where two data streams D₁ and D₂ arereproduced from the signal. D₁ and D₂ are sent through a divider 788 toa second video decoder 422 where they are converted to a video signalwhich is then delivered from an output unit 780. Also, a digital oranalogue terrestrial TV signal intercepted by a terrestrial antenna 32ais fed through an input unit 752 to the mixer 753 where one describedchannel is selected by the same manner as described in FIG. 63 anddetected to a low frequency base band signal. The signal of analogueform is sent directly to the demodulator 35 for demodulation. The signalof digital form is then fed to a discrimination/reproducing circuit 757where two data streams D₁ and D₂ are reproduced from the signal. D₁ andD₂ are converted by the second video decoder 422 to a video signal whichis then delivered further. A satellite analogue TV signal is transferredto a video demodulator 788 7880where it is AN modulatedAM demodulated toan analogue video signal which is then delivered from the output unit780. As understood, the mixer 753 of the TV receiver 781 shown in FIG.67 is arranged compatible between two, satellite and terrestrial,broadcast services. Also, a receiver circuit including a detector 755and an LPF 756 for AM modulation of an analogue signal can be utilizedcompatible with a digital ASK signal of the terrestrial TV service. Themajor part of the arrangement shown in FIG. 67 is arranged forcompatible use, thus minimizing a circuitry construction.

According to the embodiment, a 4-level ASK signal is divided into two,D₁ and D₂, level components for execution of the one-bit modemulti-level signal transmission. If an 8-level ASK signal—i.e. 8-levelVSB—is used as shown in the constellation diagram of 8-VSB signal ofFIGS. 68(a) and 68(b), it can be transmitted in a total of three bitsper symbol multi-level transmission, each bit corresponding to each ofthree-level, D₁, D₂, and D₃, arrangement. First of all, a method ofcoding the first bit will be explained. As shown in FIG. 68(a), D₁ isassigned to eight signal points 721a, 721b, 722a, 722b, 723a, 723b,724a, 724b, each pair representing a two-bit pattern. Next, for codingof the second bit, D₂ is assigned to four small signal point groups 721,722, 723, 724, each two groups representing a two-bit pattern, and D₃ isassigned to two large signal point groups 725 and 726 representing atwo-bit pattern. More particularly, this is equivalent to a form inwhich each of the four signal points 721, 722, 723, 724 shown in FIG. 57is divided into two components thus producing three different level dataat most.

The three-level signal transmission of digital HDTV signal and the likeis identical to that described in the third and fourth embodiments andwill no further be not be further explained in detail.

Here, the effect of TV broadcast using VSB of FIG. 68 will be explained.Although 8 VSB has large capacity of transmitting information, its errorrate is higher than that of 4 VSB with respect to the same C/N value.However, for the high quality HDTV broadcast, its large transmissioncapacity will allow many error correction codes, thereby lowing loweringthe error rate and assuring the future multi-level TV broadcast.

The effects of 4 VSB, 8 VSB and 16 VSB will be comparatively explained.In the case where the terrestrial broadcast service uses the frequencyband of NTSC or PAL, the substantial transmission band of approximately5 MHz will be allowed for NTSC, because NTSC is subjected to the bandlimitation of 6 MHz, as shown in FIG. 136. For 4 VSB, as its frequencyutilization efficiency is 4 bit/Hz, it will have a data transmissioncapacity equivalent to 5 MHz×4=20 Mbps. On the other hand, at least 15to 18 Mbps is required for transmission of a digital HDTV signal.Accordingly, 4 VSB is insufficient in data capacity. More specifically,as illustrated in FIG. 169, only 10 to 20% of the substantialtransmission capacity of the HDTV signal is used for the errorcorrection coding.

For 8 VSB, as its frequency utilization efficiency is 6 bit/Hz, it willhave a data transmission capacity equivalent to 5 MHz×6=30 Mbps. Asexplained above, at least 15 to 18 Mbps is required for transmission ofa digital HDTV signal. Accordingly, as illustrated in FIG. 168 169, the8 VSB modulation method can utilize, for the error correction coding,the data capacity equivalent to 50% or more of the substantialtransmission capacity of the HDTV signal. Accordingly, under thecondition that HDTV digital signals with the same data rate aretransmitted for the terrestrial broadcast service in the 6 MHz band, 8VSB is superior to 4 VSB in allowing a large number of error correctioncodes. As indicated by error rate curves 805 and 806 of FIG. 163, theTCM-8VSB having a high code gain for error correction showserror-corrected error rates smaller than those of the 4 VSB having a lowcode gain for error correction. Therefore, compared with 4 VSB, the 8VSB with high code gain for error correction can expand the service areaof the terrestrial TV broadcast. The 8 VSB is disadvantageous inincrease of error correcting circuits, which will result incomplicatedness of the circuit construction of a receiver. However, theVSB ASK system is characterized by an amplitude modulation method.Therefore, the receiver is normally equipped with an equalizer whosecircuit scale is smaller than that for the QAM method including phasecomponents. For this reason, even if the error correction circuits areadded, an overall circuit scale of the 8 VSB system will be smaller thanthat of the 32 QAM system. Hence, the 8 VSB system can provide a digitalHDTV receiver having an enlarged service area and adequate in theoverall circuit scale.

Specific example of the error correcting system will be explained later.The ECC 744a and the Trellis encoder 744b, shown in the block diagramsof a transmitter/receiver of FIGS. 84, 131, 137, 156 and 157, will beused. And, the VSB modulator 749 of 4 VSB, 8 VSB and 16 VSB explainedwith reference to FIG. 61 will be used to transmit the signal. In areceiver, the VSB demodulator 760 explained with reference to FIG. 63will be used to reproduce a digital data from 4 VSB, or 8 VSB, or 16 VSBsignal using 4, or 8, or 16-level slicer 757. Then, using the Trellisdecoder 759b and the ECC decoder 759a shown in FIGS. 84, 131, 137, 156and 157, the signal is error corrected. Thereafter, by the imageexpander of the image decoder 402, a digital HDTV signal is reproducedand output.

The ECC encoder 744a, as illustrated in FIGS. 160(a) and 160(b)explained in the sixth embodiment, comprises a Reed Solomon encoder 744jand an interleaver 744k. The ECC decoder 759a comprises a de Interleaverde-interleaver 759k and a Reed Solomon decoder 759j. As explained in theprevious embodiment, interleaving is effective to provide a systemrobust against burst errors.

To lower the error rate, the code gain can be further increased byadopting the Trellis encoder shown in FIG. 128(a), 128(b), 128(c),128(d), 128(e) and 128(f). The Trellis encoder 744b 743c and the decoder759b with ratio 2/3 will be applicable to the 8 VSB system, as shown inFIG. 172.

The above embodiments were explained based on the example wherein themulti-level digital TV signal is transmitted. Although the multi-leveldigital TV signal realizes an ideal broadcast service, the cost will notbe acceptable at the initiation of the broadcast service because ofcomplicated circuits, such as image compression circuits andmodulator/demodulator circuits. As explained in the introductory part ofthe fifth embodiment, to realize a TV broadcast service with asimplified circuit, it will be preferable to adopt the non multi-levelTV signal transmission system by equalizing the signal point distancesof 4 VSB and 8 VSB—i.e. L=L_(o), and simplifying the circuit from FIG.137 to FIG. 157. If the broadcast service is sufficiently spread,grade-up to the 8 VSB system will be considered.

Besides the combination of 4 VSB and 8 VSB, the combination of 16 VSBand 32 VSB will be explained with reference to FIGS. 159(a) to 159(d).FIG. 159(a) shows the constellation of 16 VSB. FIG. 159(b) divides thesignal points into signal groups 722a to 722h, each consisting of twosignal points, thereby realizing two-level signal transmission system byregarding these signal groups at eight signal points of 8 VSB. In thiscase, the multi-level signal transmission system will be also realizedby transmitting 8 VSB signals intermittently using the time divisionmultiplex. According to this method, the maximum data rate will bereduced to 2/3. FIG. 157(c) further divides the signals into four signalgroups 723a to 723d, thereby allowing these four signal groups to beregarded as four signal points of 4 VSB. Thus, one more level is added.In this case, the multi-level signal transmission system will be alsorealized by transmitting 4 VSB signals intermittently using the timedivision multiplex, although the maximum data rate is reduced. In thismanner, a three-level VSB is realized.

According to this multi-level data transmission system, 8 VSB or 4 VSBdata can be reproduced even if the C/N rate of 16 VSB is worsened.Furthermore, as illustrated in FIG. 159(d), 32 VSB signal transmissionwill be realized by doubling the number of the signal points of 16 VSB.If enlargement of the capacity of 16 VSB is required in the future, this32 VSB signal transmission system will provide a data capacity up to 5bits per symbol without losing the compatibility.

Above-describedThe above-described system will be embodied as a VSBreceiver shown in FIG. 161 and a VSB transmitter shown in FIG. 162.Although the explanation was based on 4 VSB and 8 VSB, the signaltransmission can be realized by using 16 VSB as shown in FIGS. 159(a),159(b) and 159(c). For 16 VSB, if used in the terrestrial broadcastservice, the signal transmission capacity of 40 Mbps is obtained in the6 MHz band. However, as the data rate of a HDTV digital compressionsignal is 15 to 18 Mbps when the MPEG regulation is adopted, the surplusof the signal transmission capacity is too much. Namely, as illustratedin FIG. 169, the redundance of R₁₆ is approximately 100% or more. Such alarge redundance is not desirable for transmitting 1-channel digitalHDTV signal. Effect brought by this 16 VSB is not so large when comparedwith the 8 VSB system, while the complicatedness of the circuitincreases alone. To provide a 2-channel HDTV terrestrial broadcast, theredundancy of 16 VSB will be reduced to the same level as 4 VSB, whichwill not allow sufficient error correcting codes to be enter entered.This means that the service area is narrowed. As described previouslythe 4 VSB system cannot provide a wide service area, because theredundance of 4 VSB is in a range of 10 to 20%. As apparent from FIG.169, the 8 VSB has the redundance of approximately 50%, which isadequately used for error correction codes. The service area can beenlarged without increasing the scale of error correction circuits somuch. Accordingly, under the condition that the digital HDTV terrestrialbroadcast service is realized in the limited band of 6 to 8 MHz, it isconcluded that the 8 VSB system is most effective and appropriate forthe VSB modulation system, as apparent from FIG. 169.

In particular, the arrangement of the video encoder 401 of the thirdembodiment shown in FIG. 30 is replaced with a modification of whichblock diagram is FIG. 69. The operation of the modified arrangement issimilar and will no longer be not be further explained in detail. Twovideo signal divider circuits 404 and 404a which may be sub-band filtersare provided forming a divider unit 794. The divider unit 794 may alsobe arranged more simple a simply as shown in the block diagram of FIG.70, in which a signal passes across one signal divider circuit two timesat time division mode. More specifically, a video signal of e.g. HDTV orsuper HDTV from the input unit 403 is time-base compressed by atime-base compressor 795 and fed to the divider circuit 404 where it isdivided into four components, H_(H)V_(H)-H, H_(H)V_(L)-H, andH_(L)V_(H)-H, and H_(L)V_(L)-H at a first cycle. At the time, fourswitches 765, 765a, 765b, 765c remain turned to the position 1 so thatH_(H)V_(H)-H, H_(H)V_(L)-H, and H_(L)V_(H)-H are transmitted to acompressing circuit 405. Meanwhile, H_(L)V_(L)-H is fed back through theterminal 1 of the switch 765c to the time-base compressor 795. At asecond cycle, the four switches 765, 765a, 765b, 765c turned to theposition 2 and all the four components of the divider circuit 404 aresimultaneously transferred to the compressing circuit 405. Accordingly,the divider unit 796 794of FIG. 70 arranged for time division processingof an input signal can be constructed in a simpler dividing circuitform.

At the receiver side, such a video decoder as described in the thirdembodiment and shown in FIG. 30 is needed for three-level transmissionof a video signal. More particularly, a third video decoder 423 isprovided which contains two mixers 556 and 556a of different processingcapability as shown in the block diagram of FIG. 71.

Also, the third video decoder 423 may be modified in which the sameaction is executed with one single mixer 556 as shown in FIG. 72. At thefirst timing, five switches 765, 765a, 765b, 765c, 765d remains remainturned to the position 1. Hence, H_(L)V_(L), H_(L)V_(H), H_(H)V_(L), andH_(H)V_(H) are fed from a first 522, a second 522a, a third 522b and afourth expander 522c to through their respective switches to the mixer556 where they are mixed to a single video signal. The video signalwhich represents H_(L)V_(L)-H of an input high resolution video signalis then fed back through the terminal 1 of the switch 765d to theterminal 2 of the switch 765c. At the second timing, the four switches765, 765a, 765b, 765c are turned to the point 2. Thus, H_(H)V_(H)-H,H_(H)V_(L)-H, H_(L)V_(H)-H, and H_(L)V_(L)-H are transferred to themixer 556 where they are mixed to a single video signal which is thensent across the terminal 2 of the switch 765d to the output unit 554 forfurther delivery.

In this manner of time division processing of a three-level signal, twomixers can be replaced with one mixer.

More particularly, four components H_(L)V_(L), H_(L)V_(H), H_(H)V_(L),H_(H)V_(H) are fed to produce H_(L)V_(L)-H at the first timing. Then,H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H are fed at the secondtiming delayed from the first timing and mixed with H_(L)V_(L)-H to atarget video signal. It is thus essential to perform the two actions atan interval of time.

If the four components are overlapped with each other or supplied in avariable sequence, they have to be time-base adjusted to a givensequence through using memories accompanied with their respectiveswitches 765, 765a, 765b, 765c. In the foregoing manner, a signal istransmitted from the transmitter at two different timing periods asshown in FIG. 73 so that no time-base controlling circuit is needed inthe receiver which is thus arranged more compact compactly.

As shown in FIG. 73, D₁ is the first data stream of a transmittingsignal and H_(L)V_(L), H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) aretransmitted on D₁ channel at the period of first timing. Then, at theperiod of second timing, H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) aretransmitted on D₂ channel. As the signal is transmitted in a timedivision sequence, the encoder in the receiver can be arranged moresimple simply.

The technique of reducing the number of the expanders in the decoderwill now be explained. FIG. 74(b) shows a time-base assignment of fourdata components 810, 810a, 810b, 810c of a signal. When other four datacomponents 811, 811a, 811b, 811c are inserted between the four datacomponents 811, 811a, 811b, 811c respectively, the latter can betransmitted at intervals of time. In action, the second video decoder422 shown in FIG. 74(a) receives the four components of the first datastream D₁ at a first input 521 and transfers them through a switch 812to an expander 503 one after another. More particularly, the component810 first fed is expanded during the feeding of the component 811 andafter completion of processing of the component 810, the succeedingcomponent 810a is fed. Hence, the expander 503 can process a row of thecomponents at time intervals by the same time division manner as of themixer, thus substituting the simultaneous action of a number ofexpanders.

FIG. 75 is a time-base assignment of data components of an HDTV signal,in which H_(L)V_(L)(1) of an HDTV component of the first channel signalfor a TV program is allocated to a data domain 821 of D₁ signal. Also,H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) carrying HDTV additionalcomponents of the first channel signal are allocated to three domains821a, 821b, 821c of D₂ signal respectively. There are provided otherdata components 822, 822a, 822b, 822c between the data components of thefirst channel signal which can thus be expanded with an expander circuitduring transmission of the other data. Hence, all the data components ofone channel signal will be processed by a single expander capable ofoperating at a higher speed.

Similar effects will be ensured by assignment of the data components toother domains 821, 821a, 821b, 821c as shown in FIG. 76. This becomesmore effective in transmission and reception of a common 4 PSK or ASKsignal having no different digital levels.

FIG. 77 shows a time-base assignment of data components during physicaltwo-level transmission of three different signal level data:e.g. NTSC,HDTV, and super HDTV or low resolution NTSC, standard resolution NTSC,and HDTV. For example, for transmission of three data components of lowresolution NTSC, standard NTSC, and HDTV, the low resolution NTSC orH_(L)V_(L) is allocated to the data domain 821 of D₁ signal. Also,H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) of the standard NTSC componentare allocated to three domains 821a, 821b, 821c respectively.H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H of the HDTV component areallocated to domains 823, 823a, and 823b respectively.

The 4 VSB or 8 VSB is associated with such a logic level arrangementbased on discrimination in the error correction capability as describedin the second embodiment as shown in FIGS. 156 and 170. Moreparticularly, H_(L)V_(L) is carried on D¹⁻¹ channel of the D₁ signal.The D₁₋₁ channel is higher in the error correction capability than D₁₋₂channel, as described in the second embodiment. The D₁₋₁ channel ishigher in the redundancy but lower in the error rate than the D₁₋₂channel and the date data 821 can be reconstructed at a lower C/N ratethan that of the other data 821a, 821b, 821c. More specifically, a lowresolution NTSC component will be reproduced at a far location from thetransmitter antenna or in a signal attenuating or shadow area, e.g. theinterior of a vehicle. In view of the error rate, the data 821 of D₁₋₁channel is less affected by signal interference than the other data821a, 821b, 821c of D₁₋₂ channel, while being specifically discriminatedand stayed in a different logic level, as described in the secondembodiment. While D₁ and D₂ are divided into two physically differentlevels, the levels determined by discrimination of the distance betweenerror correcting codes are arranged different differently in the logiclevel.

The demodulation of D₂ data requires a higher C/N rate than that for D₁data. In action, H_(L)V_(L) or low resolution NTSC signal can at leastbe reproduced in a distant or lower C/N service area. H_(L) _(V) _(H),H_(H)V_(L), and H_(H)V_(H) can in addition be reproduced at a lower C/Narea. Then, at a high C/N area, H_(L)V_(H)-H, H_(H)V_(L)-H, andH_(H)V_(H)-H components can also be reproduced to develop an HDTVsignal. Accordingly, three different level broadcast signals can beplayed back. This method allows the signal receivable area shown in FIG.53 to increase from a double region to a triple region, as shown in FIG.90, thus ensuring a higher opportunity for enjoying TV programs FIGS. .FIG. 78 is a block diagram of the third video decoder arranged for thetime-base assignment of data shown in FIG. 77, which is similar to thatshown in FIG. 72 except that the third input 551 for D₃ signal iseliminated and the arrangement shown in FIG. 74(a) is added.

In operation, both the D₁ and D₂ signals are fed through two input units521, 530 respectively to a switch 812 at the first timing. As theircomponents including H_(L)V_(L) are time divided, they are transferredin a sequence by the switch 812 to an expander 503. This sequence willnow be explained referring to the time-base assignment of FIG. 77. Acompressed form of H_(L)V_(L) of the first channel is first fed to theexpander 503 where it is expanded. Then, H_(L)V_(H), H_(H)V_(L), andH_(H)V_(H) are expanded. All the four expanded components are sentthrough a switch 812a to a mixer 554 where they are mixed to produceH_(L)V_(L)-H. H_(L)V_(L)-H is then fed back from the terminal 1 of aswitch 765a through the input 2 of a switch 765 to the H_(L)V_(L) inputof the mixer 556.

At the second timing, H_(L)V_(H)-H, H_(H)V_(L)-H, and H_(H)V_(H)-H ofthe D₂ signal shown in FIG. 77 are fed to the expander 503 where theyare expanded before being transferred through the switch 821a to themixer 556. They are mixed by the mixer 556 to an HDTV signal which isfed through the terminal 2 of the switch 765a to the output unit 521 forfurther delivery. The time-base assignment of data components fortransmission, shown in FIG. 77, contributes to the simplest arrangementof the expander and mixer. Although FIG. 77 shows two, D₁ and D₂, signallevels, four-level transmission of a TV signal will be feasible usingthe addition of a D₃ signal and a super resolution HDTV signal.

FIG. 79 illustrates a time-base assignment of data components of aphysical three-level, D₁, D₂, D₃, TV signal, in which data components ofthe same channel are so arranged as not to overlap with one another withtime. FIG. 80 is a block diagram of a modified video decoder 423,similar to FIG. 78, in which a third input 521a is added. The time-baseassignment of data components shown in FIG. 79 also contributes to thesimple construction of the decoder.

The action of the modified decoder 423 is almost identical to that shownin FIG. 78 and associated with the time-base assignment shown in FIG. 77and will no more be not be further explained. It is also possible tomultiplex data components on the D₁ signal as shown in FIG. 81. However,two data 821 and 822 are increased higher in the error correctioncapability than other data components 821a, 812b, 812c, 821b, 821c, thusstaying at a higher signal level. More particularly, the data assignmentfor transmission is made in one physical level but two logic levelrelationship. Also, each data component of the second channel isinserted between two adjacent data components of the first channel sothat serial processing can be executed at the receiver side and the sameeffects as of the time-base assignment shown in FIG. 79 will thus beobtained.

The time-base assignment of data components shown in FIG. 81 is based onthe logic level mode and can also be carried in the physical level modewhen the bit transmission rate of the two data components 821 and 822 isdecreased to ½ or ⅓ thus to lower the error rate. The physical levelarrangement is consisted of three different levels.

FIG. 82 is a block diagram of another modified video decoder 423 fordecoding of the D₁ signal time-base arranged as shown in FIG. 81, whichis simpler in construction than that shown in FIG. 80. Its action isidentical to that of the decoder shown in FIG. 80 and will be no morenot be further explained.

As understood, the time-base assignment of data components shown in FIG.81 also contributes to the similar arrangement of the expander andmixer. Also, four data components of the D₁ signal are fed at respectivetime slices to a mixer 556. Hence, the circuitry arrangement of themixer 556 or a plurality of circuit blocks such as provided in the videomixer 548 of FIG. 32 may be arranged for changing the connectiontherebetween corresponding to each data component so that they becomecompatible in time division action and thus, minimized in circuitryconstruction.

Accordingly, the receiver can be minimized in the overall construction.

It would be understood that the fifth embodiment is not limited to ASKmodulation and the other methods including PSK and QAM modulation, suchas described in the first, second, and third embodiment, will beemployed with equal success.

Also, FSK modulation will be eligible in any of the embodiments. Forexample, the signal points of a multiple-level FSK signal consisting offour frequency components f1, f2, f3, f4 are divided into groups asshown in FIG. 58 and when the distance between any two groups is suchthat the groups are spaced from each other for ease of discrimination,the multi-level transmission of the FSK signal can be implemented, asillustrated in FIG. 83.

More particularly, it is assumed that the frequency group 841 of f1 andf2 is assigned D₁=0 and the group 842 of f3 and f4 is assigned D₁=1. Iff1 and f3 represent 0 at D₂ and f2 and f4 represent 1 at D₂, two-bitdata transmission, one bit at D₁ or D₂, will be possible as shown inFIG. 83. When the C/N rate is high, a combination of D₁=0 and D₂=1 isreconstructed at t=t3 and a combination of D₁=1 and D₂=0 at t=t4. Whenthe C/N rate is low, D₁=0 only is reproduced at t=t3 and D₁=1 at t=t4.In this manner, the FSK signal can be transmitted in the multi-levelarrangement. This multi-state FSK signal transmission is applicable toeach of the third, fourth, and fifth embodiments.

The fifth embodiment may also be implemented in the form of a magneticrecord/playback apparatus of which block diagram shown in FIG. 84because its ASK mode action is appropriate to magnetic record andplayback operation.

FIG. 84 is a block diagram showing a recorder/transmitter and aplayer/receiver.

In the block diagram of FIG. 84, the VSB-ASK modulation system in thefifth embodiment comprising the transmitter 1 and the receiver 43becomes identical in constitution by replacing the transmission circuit5a of the transmitter by a recorder magnetic recording signal amplifier857a, and the reception circuit 24a of the receiver 43 by a magneticreproducing signal amplifier 857b.

Describing the operation, the HDTV signal is divided into two sets ofdata to be compared by the video encoder 401, a first data stream issubjected to error coding in the ECC encoder 743a, and a second datastream is subjected to error coding in the ECC 744a, then furthersubjected to Trellis coding in the Trellis encoder 744b to enter themodulator 749 of VSB-ASK. In the case of the recorder, after applying anoffset signal by the offset generator 856, data is recorded on amagnetic tape 855 by a recording circuit 853. In the case of thetransmitter 1, a DC offset voltage is superposed on the VSB-ASK signalby an offset voltage generator 856, and the signal is transmitted by theup converter 5a. Thus, it is easy to reproduce the synchronous signal ofthe receiver. The transmitted VSB-ASK signals of 4 VSB, 8 VSB and 16 VSBare received by the antenna 32b, and fed into a demodulator 852a by wayof a down converter 24a.

On the other hand, the signal recorded by the recorder is reproduced bya reproducing head 854a, and is sent into a demodulator 852b through areproducing circuit 858.

The input signal is demodulated by an ASK demodulator 852b such as VSBthrough a filter 858a of the demodulator 852b, and the demodulated firstdata stream is subjected to error correction by the ECC decoder 758a,and the second data row is subjected to error correction by the Trellisdecoder 759b and the ECC 759a. In consequence, the HDTV expanded tovideo signal, TV signal, or SDTV signal is delivered from the videodecoder 402. By Trellis coding, the error rate is lowered, thetransmission distance of the transmitter is extended, and the imagequality of the recording and reproducing apparatus is improved. In sucha case, the filter 858a of the receiver 43, by using a comb filter whichpossesses such a filter characteristic as to eliminate the carrier ofanalog TV signal as shown in FIG. 134, can eliminate the interference ofthe analog TV signal, and the error rate can be reduced. In this case,if the filter is always placed, the signal deteriorates. To avoid this,as shown in FIG. 65, the analog TV filter 760a is turned on only whenthe signal deteriorates due to interference of the analog TV by theerror rate detector 782, and is turned off when there is nointerference, so that deterioration by filter can be prevented.

In the case shown in FIG. 84, of the first data stream and second datastream, the error rate is smaller in the second data stream. Therefore,by transmitting/recording high priority (HP) information such asde-scramble information, and header information of image data in FIG. 66into the second data stream, image reproduction of de-scramble or eachblock of image can be stabilized.

As shown in FIGS. 137 and 172, high priority information (HP) istransmitted by the subchannel with a higher code gain by changing thecode gain of Trellis decoder or ECC decoder error correction by therespective subchannel for data row time shared by the transmitter of 8VSB and 16 VSB. Since the error rate of HP information decreases, evenif noise is generated to a certain degree in the transmission route todeteriorate signals which results in destruction of Low priorityinformation(LP), HP information data will remain undestructed. Bytransmitting descramble information and header information such as datapacket address of each image block as HP information, descramblestabilizes for a long time and viewers can watch descrambled programs.At the same time, since destruction of each image block is prevented,viewers can watch TV programs with acceptable image quality as imagequality deteriorates only generally even when received signalsdeteriorate.

Embodiment 6

The communication and/or recording method of the present invention isapplicable to a magnetic recording and playback apparatus. Although thepresent invention is applied for a multiple-level ASK data transmissionin the above-described fifth embodiment, it is also feasible in the samemanner to adopt this invention in a magnetic recording and playbackapparatus of a multi-level ASK recording system, as shown in FIG. 173. Amulti-level or non multi-level magnetic recording can be realized byapplying the C-CDM method of the present invention to PSK, FCK FSK, andQAM, as well as ASK.

First of all, the method of realizing a multi-level recording in a 16QAM or 32 QAM magnetic recording playback apparatus will be explainedusing an example of the C-CDM method of the present invention. FIG. 84is a circuit block diagram wherein the C-CDM is applied to 16 QAM, 32QAM, 4ASK 4 ASK, 8 ASK, 16 ASK and 8 PSK. Hereinafter, a QAM systembeing multiplexed by the C-CDM method is termed as SRQAM. FIGS. 137 and154 show block diagrams applicable for the signal transmission system ofthe broadcast.

As shown in FIG. 84, an input video signal, e.g. an HDTV signal, to amagnetic record/playback apparatus 851 is divided and compressed by avideo encoder 401 into a low frequency band signal through a first videoencoder 401a and a high frequency band signal through a second videoencoder 401b respectively. Then, a low frequency band component, e.g.H_(L)V_(L), of the video signal is fed to a first data stream input 743of an input unit 742 and a high frequency band component includingH_(H)V_(H) is fed to a second data stream input 744 of the same. The twocomponents are further transferred to a modulator 749 of amodulator/demodulator unit 852 852a. The first data stream input 743adds an error correcting code to the low frequency band signal in an ECC743a. On the other hand, the second data stream fed into the second datastream input 744 is 2 bit in case of 16 SRQAM, 3 bit in case of 36SRQAM, and 4 bit in case of 64 SRQAM. After an error correcting codebeing encoded by an ECC 744a, this signal is supplied to a Trellisencoder 744h, or 744g, or 744b shown in FIGS. 128(a), 128(b) and 128(c)in which a Trellis encoded signal having a ratio 1/2 for 16 SRQAM, 2/3for 32 SRQAM, and 3/4 for 64 SRQAM is produced. A 64 SRQAM signal, forexample, has a first data stream of 2 bit and a second data stream of 4bit. A Trellis encoder 744b of FIG. 128(c) allows this 64 SRQAM signalto perform a Trellis encoding of ratio 3/4 wherein 3 bit data isconverted into 4 bit data. For 4 ASK, 8 ASK and 16 ASK, the Trellisencoding of 1/2, 2/3 and 3/4 is independently performed. Thus redundancyincreases and a data rate decreases, while error correcting capabilityincreases. This results in the reduction of an error rate in the samedata rate. Accordingly, transmittable information amount of therecording/playback system or transmission system will increasesubstantially. As the 8 VSB transmission system explained in the fifthembodiment is three bits per symbol, the Trellis encoder 744g and theTrellis decoder 744q of ratio 2/3, shown in FIGS. 128(b) and 128(e) canbe used for this 8 VSB transmission system. An overall block diagram isshown in FIG. 171.

It is however, possible to constitute the first data stream input 743not to include a Trellis encoder as shown in FIG. 84 of this sixthembodiment because the first data stream has low error rate inherently.This will be advantageous in view of simplification of circuitconfiguration. The second data stream, however, has a narrow inter-codedistance as compared with the first data stream and, therefore, has aworse error rate. The Trellis encoding of the second data streamimproves such a worse error rate. There is no doubt that an overallcircuit configuration becomes simple if the Trellis encoding of thefirst data stream is eliminated. An operation for modulation is almostidentical to that of the transmitter of the fifth embodiment shown inFIG. 64 and will not be further explained. A modulated signal of themodulator 749 is fed into a recording/playback circuit 853 in which itis AC biased by a bias generator 856 and amplified by an amplifier 857a.Thereafter, the signal is fed to a magnetic head 854 854b for recordingonto a magnetic tape 855.

A format of the recorded signal is shown in a recording signal frequencyassignment of FIG. 113. A main, e.g. 16 SRQAM, signal 859 having acarrier of frequency fc records information, and also a pilot f_(p)signal 859a having a frequency 2fc is recorded simultaneously.Distortion in the recording operation is lowered as a bias signal 859bhaving a frequency f_(BIAS) adds AC bias for magnetic recording. Two ofthree-level signals shown in FIG. 113 are recorded in multiple state. Inorder to reproduce these recorded signals, two thresholds Th-1-2, Th-2are given. A signal 859 will reproduce all of two levels while a signal859c will reproduce D₁ data only, depending on C/N level of therecording/playback.

A main signal of 16 SRQAM will have a signal point assignment shown inFIG. 10. Furthermore, a main signal of 36 SRQAM will have a signal pointassignment shown in FIG. 100. Furthermore, a main signal of 4 ASK or 8ASK will have a signal point assignment of FIG. 58 or FIG. 68(a) and68(b). In reproduction of this signal, both the main signal 859 and thepilot signal 859a are reproduced through the magnetic head 854 854a andamplified by an amplifier 857b. An output signal of the amplifier 857bis fed to a carrier reproduction circuit 858 in which a filter 858aseparates the frequency of the pilot signal fp having a frequency 2f02f₀ and a ½ frequency divider 858b reproduces a carrier of frequencyf0f₀ to transfer it to a demodulator 760. This reproduced carrier isused to demodulate the main signal in the demodulator 760. Assuming thata magnetic recording tape 855, e.g. HDTV tape, is of a high C/N rate, 16signal points are discriminated and thus both D₁ and D₂ are demodulatedin the demodulator 760. Subsequently, a video decoder 402 reproduces allthe signals. An HDTV VCR can reproduce a high bit-rate TV signal such as15 Mbps HDTV signal. The lower the C/N rate is, the cheaper the cost ofa video tape is. So far, a VHS tape in the market is inferior by morethan 10 dB in C/N rate to a full-scale broadcast tape. If a video tape855 is of low C/N rate, it will not be able to discriminate all the 16or 32 valued signal points. Therefore the first data stream D₁ can bereproduced, while a 2 bit, 3 bit, or 4 bit data stream of the seconddata stream D₂ cannot be reproduced. Only 2 bit data stream of the firstdata stream is reproduced. If a two-level HDTV video signal is recordedand reproduced, a low C/N tape having insufficient capability ofreproducing a high frequency band video signal can output only a lowrate low frequency band video signal of the first data stream,specifically e.g. a 7 Mbps wide NTSC TV signal.

As shown in a block diagram of FIG. 114, a second data stream output759, the second data stream input 744, and the second video decoder 402acan be eliminated in order to provide customers one aspect of lowergrade products. In this case, a recording/playback apparatus 851,dedicated to a low bit rate, will include a modulator such as amodulated QPSK which modulates or demodulates the first data streamonly. This apparatus allows only the first data stream to be recordedand reproduced. Specifically, a wide NTSC grade video signal can berecorded and reproduced.

Above-described high C/N rate video tape 855 capable of recording a highbit-rate signal, e.g. HDTV signal, will be capable of being used in sucha low bit-rate dedicated magnetic recording/playback apparatus but willreproduce the first data stream D₁ only. That is, the wide NTSC signalis outputted, while the second data stream is not reproduced. In otherwords, one recording/playback apparatus having a complicatedconfiguration can reproduce a an HDTV signal and the otherrecording/playback apparatus having a simple configuration can reproducea wide NTSC signal if a given video tape 855 includes the samemulti-level HDTV signal. Accordingly in case of two-level multiplestate, four combinations will be realized with perfect compatibilityamong two tapes having different C/N rates and two recording/playbackapparatus having different recording/playback data rates. This willbring remarkable effect. In this case, an NTSC dedicated apparatus willbe simple in construction as compared with an HDTV dedicated apparatus.In more detail, a circuitry scale of EDTV decoder will be ⅙ of that ofHDTV decoder. Therefore, a low function apparatus can be realized atfairly low cost. Realization of two, HDTV and EDTV, typesrecording/playback apparatus having different recording/reproducingcapability of picture quality will provide various type products rangingin a wide price range. Users can freely select a tape among a pluralityof tapes from an expensive high C/N rate tape to a cheaper low C/N ratetape, as occasion demands so as to satisfy required picture quality. Notonly maintaining perfect compatibility but obtaining expandablecapability will be attained and further compatibility with a futuresystem will be ensured. Consequently, it will be possible to establishlong-lasting standards for recording/playback apparatus. Other recordingmethods will be used in the same manner. For example, a multi-levelrecording will be realized by use of phase modulation explained in thefirst and third embodiments. A recording by the ASK explained in thefifth embodiment will be also possible. A multiple state of two- orthree-layer will be realized by converting present recording fromtwo-level to four-level ASK or eight-level ASK and dividing into twogroups as shown in FIGS. 59(c), 59(d), 68(a) and 68(b).

A circuit block diagram for ASK is shown in FIG. 173, which is identicalto that disclosed in FIG. 84. With the combination of the Trellis andASK, the error rate can be lowered. Besides embodiments alreadydescribed, a multi-level recording will be also realized by use ofmultiple tracks on a magnetic tape. Furthermore, a theoreticalmulti-level recording will be feasible by differentiating the errorcorrecting capability so as to discriminate respective data.

Compatibility with future standards will be described below. A settingof standards for recording/playback apparatus such as VCR is normallydone by taking account of the most highest C/N rate tape available inpractice. The recording characteristics of a tape progresses rapidly.For example, the C/N rate has been improved more than 10 dB comparedwith the tape used 10 years ago. If supposed that new standards will beestablished after 10 to 20 years due to an advancement of tape property,a conventional method will encounter with the difficulty in maintainingcompatibility with older standards. New and old standards, in fact, usedto be one-way compatible or non-compatible with each other.

On the contrary, in accordance with the present invention, the standardsare first of all established for recording and/or reproducing the firstdata stream and/or second data stream on present day tapes.Subsequently, if the C/N rate is improved magnificently in future, anupper level data stream, e.g. a third data stream, will be added withoutany difficulty as long as the present invention is incorporated in thesystem. For example, a super HDTV VCR capable of recording orreproducing three-level 64 SRQAM or 8 ASK will be realized whilemaintaining perfect compatibility with the conventional standards. Amagnetic tape, recording first to third data streams in compliance withnew standards, will be able to use, of course, in the older two-levelmagnetic recording/playback apparatus capable of recording and/orreproducing only first and second data streams. In this case, first andsecond data streams can be reproduced perfectly although the third datastream is left non-reproduced. Therefore, an HDTV signal can bereproduced. For these reasons, the merit of expanding recording dataamount while maintaining compatibility between new and old standards isexpected.

Returning to the explanation of reproducing operation of FIG. 84, themagnetic head 854 854a and the magnetic reproduction circuit 853858reproduce a reproducing signal from the magnetic tape 855 and feedsfeed it to the modulation/demodulation circuit 852demodulator unit 852b.The demodulating operation is almost identical with that of first,third, and fourth embodiments and will no further be not be furtherexplained. The demodulator 760 reproduces the first and second datastreams D₁ and D₂. The second data stream D₂ is error corrected withhigh code gain in a Trellis-decoder 759b such as a Vitabi decoder, so asto be low error rate. The video decoder 402 demodulates D₁ and D₂signals to output an HDTV video signal.

FIG. 131 is a block diagram showing a three-level magneticrecording/playback apparatus in accordance with the present inventionwhich includes one theoretical level in addition to two physical levels.This system is substantially the same as that of FIG. 84. The differenceis that the first data stream is further divided into two subchannels byuse of a TDM in order to realize a three-level constitution.

As shown in FIG. 131, an HDTV signal is separated first of all into two,medium and low frequency band video signals D₁₋₁, and D₁₋₂, through a1-1 video encoder 401c and a 1-2 video encoder 401d and, thereafter, fedinto a first data stream input 743 of an input section 742. The datastream D₁₋₁ having a picture quality of MPEG grade is error correctingcoded with high code gain in an ECC coder 743a, while the data streamD₁₋₂ is error correcting coded with normal code gain in an ECC encoder743b, D₁₋₁, and D₁₋₂ are time multiplexed together in a TDM 743c to beone data stream D1. D₁ and D₂ are modulated into two-level signals in aC-CDM 749 and then recorded on the magnetic tape 855 through themagnetic head 854.

In playback operation, a recording signal reproduced through themagnetic head 854 is demodulated into D₁ and D₂ by the C-CDM demodulator760 in the same manner as in the explanation of FIG. 84. The first datastream D₁ is demodulated into two, D₁₋₁ and D₁₋₂, subchannels throughthe TDM 758c provided in the first data stream output 758. D₁₋₁ data iserror corrected in an ECC decoder 758a having high code gain. Therefore,D₁₋₁ data can be demodulated at a lower C/N rate as compared with D₁₋₂data. A 1-1 video decoder 402a 402c decodes the D₁₋₁ data and outputs anLDTV signal. On the other hand, D₁₋₂ data is error corrected in an ECCdecoder 758b having normal code gain. Therefore, D₁₋₂ data has athreshold value of high C/N rate compared with D₁₋₁ data and thus willnot be demodulated when a signal level is not large. D₁₋₂ data is thendemodulated in a 1-2 video decoder 402d and summed with D₁₋₁ data tooutput an EDTV signal of wide NTSC grade.

The second data stream D₂ is Vitabi demodulated in a Trellis decoder759b and error corrected at an ECC decoder 759a. Thereafter, D₂ data isconverted into a high frequency band video signal through a second videodecoder 402b and, then, summed with D₁₋₁ and D₁₋₂ data to output an HDTVsignal. In this case, a threshold value of the C/N rate of D₂ data isset larger than that of C/N rate of D₁₋₂ data. Accordingly, D₁₋₁ data,i.e. an LDTV signal, will be reproduced from a tape 855 having a smallerC/N rate D₁₋₁, and D₁₋₂ data, i.e. an EDTV signal, will be reproducedfrom a tape 855 having a normal C/N rate. And, D₁₋₁, D₁₋₂, and D₂, i.e.an HDTV signal, will be reproduced from a tape 855 having a high C/Nrate.

Three-level magnetic recording/playback apparatus can be realized inthis manner. As described in the foregoing description, the tape 855 hasan interrelation between C/N rate and cost. The present invention allowsusers to select a grade of tape in accordance with a content of TVprogram they want to record because video signals having picturequalities of three grades are recorded and/or reproduced in accordancewith tape cost.

Next, an effect of multi-level recording will be described with respectto fast feed playback. As shown in a recording track diagram of FIG.132, a recording track 855a having an azimuth angle A and a recordingtrack 855b having an opposite azimuth angle B are alternately arrayed onthe magnetic tape 855. The recording track 855a has a recording region855c at its central portion and the remainder as D₁₋₂ recording regions855d, as denoted in the drawing. This unique recording pattern isprovided on at least one of several recording tracks. The recordingregion 855c records one frame of LDTV signal. A high frequency bandsignal D₂ is recorded on a D₂ recording region 855e corresponding to anentire recording region of the recording track 855a. This recordingformat causes no novel effect against a normal speed recording/playbackoperation.

A fast feed reproduction in a reverse direction does not allow amagnetic head trace 855f having an azimuth angle A to coincide with themagnetic track as shown in the drawing. As the present inventionprovides the D₁₋₁ recording region 855c at a central narrow region ofthe magnetic tape as shown in FIG. 132, this region only is surelyreproduced although it occurs at a predetermined probability. Thusreproduced D₁₋₁ signal can demodulate an entire picture plane of thesame time although its picture quality is an LDTV of MPEG 1 level. Inthis manner several to several tens LDTV signals per second can bereproduced with perfect picture images during the fast feed playbackoperation, thereby enabling users to surely confirm picture imagesduring the fast feed operation.

A head trace 855g corresponds to a head trace in the reverse playbackoperation, from which it is understood only a part of the magnetic trackis traced in the reverse playback operation. The recording/playbackformat shown in FIG. 132 however allows, even in such a reverse playbackoperation, to reproduce D₁₋₁ recording region and, therefore, ananimation of LDTV grade is outputted intermittently.

Accordingly, the present invention makes it possible to record a pictureimage of LDTV grade within a narrow region on the recording track, whichresults in intermittent reproduction of almost perfect still pictureswith picture quality of LDTV grade during normal and reverse fast feedplayback operations. Thus, the users can easily confirm picture imageseven in high-speed searching.

Next, another method will be described to respond a higher speed fastfeed playback operation. A D₁₋₁ recording region 855c is provided asshown at lower right of FIG. 132, so that one frame of LDTV signal isrecorded thereon. Furthermore, a narrow D₁₋₁· , D₂ recording region 855his provided at a part of the D₁₋₁ recording region 855c. A subchannelD₁₋₁ in this region records a part of information relating to the oneframe of LDTV signal. The remainder of the LDTV information is recordedon the D₂ recording region 855j of the D₁· , D₂ recording region 855h ina duplicated manner. The subchannel D₂ has a data recording capacity 3to 5 times as much as the subchannel D₁₋₁. Therefore, subchannels D₁₋₁and D₂ can record one frame information of LDTV signal on a smaller,⅓^(˜){fraction (1/5)}, area of the recording tape. As the head trace canbe recorded in a further narrower regions 855h, 855j, both time and areaare decreased into ⅓^(˜){fraction (1/5)} as compared with a head tracetime T_(S1). Even if the trace of head is further inclined by increasingfast feed speed amount, the probability of entirely tracing this regionwill be increased. Accordingly, perfect LDTV picture images will beintermittently reproduced even if the fast feed speed is increased up to3 to 5 times as fast as the case of the subchannel D₁₋₁ only.

In case of a two-level VCR, this method is useless in reproducing the D₂recording region 855j and therefore this region will not be reproducedin a high-speed fast feed playback operation. On the other hand, athree-level high performance VCR will allow users to confirm a pictureimage even if a fast feed playback operation is executed at a faster, 3to 5 times as fast as two-level VCR, speed. In other words, not onlyexcellent picture quality is obtained in accordance with the cost but amaximum fast feed speed capable of reproducing picture images can beincreased in accordance with the cost.

Although this embodiment utilizes a multi-level modulation system, it isneedless to say that a normal, e.g. 16 QAM, modulation system can alsobe adopted to realize the fast feed playback operation in accordancewith the present invention as long as an encoding of picture images isof multiple type.

A recording method of a conventional non-multiple digital VCR, in whichpicture images are highly compressed, disperses video data uniformly.Therefore, it was not possible in a fast feed playback operation toreproduce all the picture images on a picture plane of the same time.The picture reproduced was the one consisting of a plurality of pictureimage blocks having non-coincided time bases with each other. Thepresent invention, however, provides a multi-level HDTV VCR which canreproduce picture image blocks having coincided time bases on a pictureplane during a fast feed playback operation although its picture qualityis of LDTV grade.

The three-level recording in accordance with the present invention willbe able to reproduce a high resolution TV signal such as HDTV signalwhen the recording/playback system has a high C/N rate. Meanwhile, a TVsignal of EDTV grade, e.g. a wide NTSC signal, or a TV signal of LDTVgrade, e.g. a low resolution NTSC signal, will be outputted when therecording/playback system has a low C/N rate or poor function.

As it is described in the foregoing description, the magneticrecording/playback apparatus in accordance with the present inventioncan reproduce picture images consisting of the same content even if C/Nrate is low or error rate is high, although the resolution or thepicture quality is relatively low.

Embodiment 7

A seventh embodiment of the present invention will be described forexecution of four-level video signal transmission. A combination of thefour-level signal transmission and the four-level video dataconstruction will create a four-level. Signal level signal service areaas shown in FIG. 91. The four-level service area is consisted of, frominnermost, a first 890a, a second 890b, a third 890c, and a fourthsignal receiving area 890d. The method of developing such a four-levelservice area will be explained in more detail.

The four-level arrangement can be implemented by using four physicallydifferent levels determined through modulation or four logic levelsdefined by data discrimination in the error correction capability. Theformer provides a large difference in the C/N rate between two adjacentlevels and the C/N rate has to be increased to discriminate all the fourlevels from each other. The latter is based on the action ofdemodulation, and a difference in the C/N rate between two adjacentlevels should stay at minimum. Hence, the four-level arrangement is bestconstructed using a combination of two physical levels and two logiclevels. The division of a video signal into four signal levels will beexplained.

FIG. 93 is a block diagram of a divider circuit 3 which comprises avideo divider 895 and four compressors 405a, 405b, 405c, 405d. The videodivider 895 contains three dividers 404a, 404b, 404c which are arrangedidentical to the divider circuit 404 of the first video encoder 401shown in FIG. 30 and will be no more not be further explained. An inputvideo signal is divided by the dividers into four components, H_(L)V_(L)or low resolution data, H_(H)V_(H) of high resolution data, andH_(L)V_(H) and H_(H)V_(L) for medium resolution data. The resolution ofH_(L)V_(L) is a half that of the original input signal.

The input video signal is first divided by the divider 404a into two,high and low, frequency band components, each component being dividedinto two, horizontal and vertical, segments. The intermediate betweenthe high and low frequency ranges is a dividing point according to theembodiment. Hence, if the input video signal is an HDTV signal of1000-line vertical resolution, H_(L)V_(L) has a vertical resolution of500 lines and a horizontal resolution of a half value.

Each of two, horizontal and vertical, data of the low frequencycomponent H_(L)V_(L) is further divided by the divider 404c into twofrequency band segments. Hence, an H_(L)V_(L) segment output is 250lines in the vertical resolution and ¼ of the original horizontalresolution. This output of the divider 404c which is termed as an LLsignal is then compressed by the compressor 405a to a D₁₋₁ signal.

The other three higher frequency segments of H_(L)V_(L) are mixed by amixer 772c to an LH signal which is then compressed by the compressor405b to a D₁₋₂ signal. The compressor 405b may be replaced with threecompressors provided between the divider 404c and the mixer 772c.

H_(L)V_(H), H_(H)V_(L), and H_(H)V_(H) formfrom the divider 404a aremixed by a mixer 772a to an H_(H)V_(H)-H signal. If the input signal isas nigh as 1000 lines in both horizontal and vertical resolution,H_(H)V_(H)-H has 500 to 1000 lines of a horizontal and a verticalresolution. H_(H)V_(H)-H is fed to the divider 404b where it is dividedagain into four components.

Similarly, H_(L)V_(L) from the divider 404b has 500 to 750 lines of ahorizontal and a vertical resolution and transferred as an HL signal tothe compressor 405c. The other three components, H_(L)V_(H), H_(H)V_(L),and H_(H)V_(H), from the divider 404b have 750 to 1000 lines of ahorizontal and a vertical resolution and are mixed by a mixer 772b to anHH signal which is then compressed by the compressor 405d and deliveredas a D₂₀₂D₂₋₂ signal after compression, the HL signal is delivered as aD₂₋₁ signal. As the result, LL or D₁₋₁ carries a frequency data of 0 to250 lines, LH or D₁₋₂ carries a frequency data from more than 250 linesup to 500 lines, HL or D₁₋₂ carries a frequency data of more than 500lines up to 750 lines, and HH or D₂₋₂ carries a frequency data of morethan 750 lines to 1000 lines so that the divider circuit 3 can provide afour-level signal. Accordingly, when the divider circuit 3 of thetransmitter 1 shown in FIG. 87 is replaced with the divider circuit ofFIG. 93, the transmission of a four-level signal will be implemented.

The combination of multi-level data and multi-level transmission allowsa video signal to be at steps declined in the picture quality inproportion to the C/N rate during transmission, thus contributing to theenlargement of the TV broadcast service area. At the receiving side, theaction of demodulation and reconstruction is identical to that of thesecond receiver of the second embodiment shown in FIG. 88 and will be nomore not be further explained. In particular, the mixer 37 is modifiedfor video signal transmission rather than data communications and willnow be explained in more detail.

As described in the second embodiment, a received signal after beingdemodulated and error corrected, is fed as a set of four componentsD₁₋₁, D₁₋₂, D₂₋₁, D₂₋₂ to the mixer 37 of the second receiver 33 of FIG.88.

FIG. 94 is a block diagram of a modified mixer 33 in which D₁₋₁, D₁₋₂,D₂₋₁, D₂₋₂ are explained by their respective expanders 523a, 523b, 523c,523d to an LL, and LH, an HL, and an HH signal respectively which areequivalent to those described with FIG. 93. If the bandwidth of theinput signal is 1, LL has a bandwidth of ¼, LL+LH has a bandwidth of ½,LL+LH+HL has a bandwidth of ¾, and LL+LH+HL+HH has a bandwidth of 1. TheLH signal is then divided by a divider 531a and mixed by a video mixer548a with the LL signal. An output of the video mixer 548a istransferred to an H_(L)V_(L) terminal of a video mixer 548c. The videomixer 531a is identical to that of the second decoder 527 of FIG. 32 andwill be no more not be further explained. Also, the HH signal is dividedby a divider 531b and fed to a video mixer 548b. At the video mixer548b, the HH signal is mixed with the HL signal to an H_(H)V_(H)-Hsignal which is then divided by a divider 531c and sent to the videomixer 548c. At the video mixer 548c, H_(H)V_(H)-H is combined with thesum signal of LH and LL to a video output. The video output of the mixer33 is then transferred to the output unit 36 of the second receivershown in FIG. 88 where it is converted to a TV signal for delivery. Ifthe original signal has 1050 lines of vertical resolution or is an HDTVsignal of about 1000-line resolution, its four different signal levelcomponents can be intercepted in their respective signal receiving areasshown in FIG. 91.

The picture quality of the four different components will be describedin more detail. The illustration of FIG. 92 represents a combination ofFIGS. 86 and 91. As apparent, when the C/N rate increases, the overallsignal level of amount of data is increased from 862d to 862a by stepsof four signal levels D₁₋₁, D₁₋₂, D₂₋₁, D₂₋₂.

Also, as shown in FIG. 95, the four different level components LL, LH,HL, and HH are accumulated in proportion to the C/N rate. Morespecifically, the quality of a reproduced picture will be increased asthe distance from a transmitter antenna becomes small. When L=Ld, LLcomponent is reproduced. When L=Lc, LL+LH signal is reproduced. WhenL=Lb, LL+LH+HL signal is reproduced. When L=La, LL+LH+HL+HH signal isreproduced. As the result, if the bandwidth of the original signal is 1,the picture quality is enhanced at ¼ increments of bandwidth from ¼ to 1depending on the receiving area. If the original signal is an HDTV of1000-line vertical resolution, a reproduced TV signal is 250, 500, 750,and 1000 lines in the resolution at their respective receiving areas.The picture quality will thus be varied at steps depending on the levelof a signal. FIG. 96 shows the signal propagation of a conventionaldigital HDTV signal transmission system, in which no signal reproductionwill be possible when the C/N rate is less than V0. Also, signalinterception will hardly be guaranteed at signal interference regions,shadow regions, and other signal attenuating regions, denoted by thesymbol x, of the service area. FIG. 97 shows the signal propagation ofan HDTV signal transmission system of the present invention. As shown,the picture quality will be a full 1000-line grade at the distance Lawhere C/N=a, a 750-line grade at the distance Lb where CM=b, a 500-linegrade at the distance Lc where C/N=c, and a 250-line grade at thedistance Ld where C/N=d. Within the distance La, there are shownunfavorable regions where the C/N rate drops sharply and no HDTV qualitypicture will be reproduced. As understood, a lower picture qualitysignal can however be intercepted and reproduced according to themulti-level signal transmission system of the present invention. Forexample, the picture quality will be a 750-line grade at the point B ina building shadow area, a 250-line grade at the point D in a runningtrain, a 750-line grade at the point F in a ghost developing area, a250-line grade at the point G in a running car, a 250-line grade at thepoint L in a neighbor signal interference area. As set forth above, thesignal transmission system of the present invention allows a TV signalto be successfully received at a grade in the area where theconventional system is poorly qualified, thus increasing its servicearea. FIG. 98 shows an example of simultaneous broadcasting of fourdifferent TV programs, in which three quality programs C, B, A aretransmitted on their respective channels D₁₋₂, D₂₋₁, D₂₋₂ while aprogram D identical to that of a local analogue TV station is propagatedon the D₁₋₁ channel. Accordingly, while the program D is kept availableat simulcast service, the other three programs can also be distributedon air for offering a multiple program broadcast service.

Embodiment 8

Hereinafter, an eighth embodiment of the present invention will beexplained referring to the drawings. The eighth embodiment employs amulti-level signal transmission system of the present invention for atransmitter/receiver in a cellular telephone system.

FIG. 115 is a block diagram showing a transmitter/receiver of a portabletelephone, in which a telephone conversation sound inputted across amicrophone 762 is compressed and coded in a compressor 405 intomulti-level, D₁₋₁, D₂, and D₃ data previously described. These D₁₋₁, D₂,and D₃ data are time divided in a time division circuit 765 intopredetermined time slots and, then, modulated in a modulator 4 into amulti-level, e.g. SRQAM, signal previously described. Thereafter, anantenna sharing unit 764 and an antenna 22 transmit a carrier wavecarrying a modulated signal, which will be intercepted by a base stationlater described and further transmitted to other base stations or acentral telephone exchanger so as to communicate with other telephones.

On the contrary, the antenna 22 receives transmission radio waves fromother base stations as communication signals from other telephones. Areceived signal is demodulated in a multiple-level, e.g. SRQAM, typedemodulator 45 into D₁₋₁, D₂, and D₃ data. A timing circuit 767 detectstiming signals on the basis of demodulated signals. These timing signalsare fed into the time division circuit 765. Demodulated signals D₁₋₁,D₂, and D₃ are fed into an expander 503 and expanded into a soundsignal, which are transmitted to a speaker 763 and converted into sound.

FIG. 116 shows a block diagram exemplarily showing an arrangement ofbase stations, in which three base stations 771, 772, and 773locatearelocated at center of respective receiving cells 768, 769, and 770 ofhexagon or circle. These base stations 771, 772, and 773 respectivelyhas have a plurality of transmitter/receiver units 761a^(˜) 761j eachsimilar to that of FIG. 115 so as to have data communication channelsequivalent to the number of these transmitter/receiver units. A basestation controller 774 is connected to all the base stations and alwaysmonitors a communication traffic amount of each base station. Based onthe monitoring result, the base station controller 774 carries out anoverall system control including allocation of channel frequencies torespective base stations or control of receiving cells of respectivebase stations.

FIG. 117 is a view showing a traffic distribution of communicationamount in a conventional, e.g. QPSK, system. A diagram d=A shows data774a and 774b having frequency utilization efficiency 2 bit/Hz, and adiagram d=B shows data 774c of frequency utilization efficiency 2bit/Hz. A summation of these data 774a, 774b, and 774c becomes a data774d, which represents a transmission amount of Ach consisting ofreceiving cells 768 and 770. Frequency utilization efficiency of 2bit/Hz is uniformly distributed. However, density of population in anactual urban area is locally high in several crowded areas 775a, 775b,and 775c which includes concentrated buildings concentrated . A data774e representing a communication traffic amount shows several peaks atlocations just corresponding to these crowded areas 775a, 775b, and775c, in contrast with other area having small communication amount. Acapacity of a conventional cellular telephone was uniformly set to 2bit/Hz frequency efficiency at entire region as shown by the data 774dirrespective of actual traffic amount TF shown by the data 774e. It isnot effective to give the same frequency efficiency regardless of actualtraffic amount. In order to compensate for this ineffectiveness, theconventional systems have allocated many frequencies to the regionshaving a large traffic amount, increased channel number, or decreasedthe receiving cell of the same. However, an increase of channel numberis restricted by the frequency spectrum. Furthermore, conventionalmulti-level, e.g. 16 QAM or 64 QAM, mode transmission systems increasetransmission power. A reduction of receiving cell will induce anincrease in number of base stations, thus increasing installation cost.

It is ideal for the improvement of an overall system efficiency toincrease the frequency efficiency of the region having a larger trafficamount and decrease the frequency efficiency of the region having asmaller traffic amount. A multi-level signal transmission system inaccordance with the present invention realizes this ideal modification.This will be explained with reference to FIG. 118 showing acommunication amount & and traffic distribution in accordance with theeighth embodiment of the present invention.

More specifically, FIG. 118 shows communication amounts of respectivereceiving cells 770b, 768, 769, 770, and 770a taken along a line A-A′.The receiving cells 768 and 770 utilize frequencies of a channel groupA, while the receiving cells 770b, 769, and 770a utilize frequencies ofa channel group B which does not overlap with the channel group A. Thebase station controller 774 shown in FIG. 116 increases or decreaseschannel number of these channels in accordance with the traffic amountof respective receiving cells. In FIG. 118, a diagram d=A represents adistribution of a communication amount of the A channel. A diagram d=Brepresents a distribution of a communication amount of the B channel. Adiagram d=A+B represents a distribution of a communication amount of allthe channels. A diagram TF represents a communication traffic amount,and a diagram P shows a distribution of buildings and population.

The receiving cells 768, 769, and 770 employ the multi-level, e.g.SRQAM, signal transmission system, Therefore, it is possible to obtain afrequency utilization efficiency of 6 bit/Hz, three times as large as 2bit/Hz of QPSK, in the vicinity of the base stations as denoted by data776a, 776b, and 776c. Meanwhile, the frequency utilization efficiencydecreases at steps from 6 bit/Hz to 4 bit/Hz, and 4 bit/Hz to 2 bit/Hz,as it goes to suburban area. If the transmission power is insufficient,2 bit/Hz areas become narrower than the receiving cells, denoted bydotted lines 777a, 777b, 777c, of QPSK. However, an equivalent receivingcell will be easily obtained by slightly increasing the transmissionpower of the base stations.

Transmitting/receiving operation of a mobile station capable ofresponding to a 64 SRQAM signal is carried out by use of modified QPSK,which is obtained by set setting a shift amount of SRQAM to S=1, at theplace far from the base station, by use of 16 SRQ M at a place not sofar from the same, and 64 SRQAM at the near place. Accordingly, themaximum transmission power does not increase as compared with QPSK.

Furthermore, 4 SRQAM type transmitter/receiver, whose circuitconfiguration is simplified as shown in a block diagram of FIG. 121,will be able to communicate with other telephones while maintainingcompatibility. That will be the same in 16 SRQAM typetransmitter/receiver shown in a block diagram of FIG. 122. As a result,three different type telephones having different modulation systems willbe provided. Small in size and light in weight is important for portabletelephones. In this regard, the 4 SRQAM system having a simple circuitconfiguration will be suitable for the users who want a small and lighttelephone although its frequency utilization efficiency is low andtherefore cost of a call may increase. In this manner, the presentinvention system can suit for a wide variety of usage.

As is explained above, the transmission system having a distributionlike d=A+B of FIG. 118, whose capacity is locally altered, isaccomplished. Therefore, an overall frequency utilization efficiencywill be much effectively improved if the layout of base stations isdetermined to fit for the actual traffic amount denoted by TF.Especially, effect of the present invention will be large in a microcell system, whose receiving cells are smaller and therefore numeroussub base stations are required. Because a large number of sub basestations can be easily installed at the place having a large trafficamount.

Next, data assignment of each time slot will be explained referring toFIG. 119, wherein FIG. 119(a) shows a conventional time slot and FIG.119(b) shows a time slot according to the eighth embodiment. Theconventional system performs a down transmission, i.e. from a basestation to a terminal station, transmission as shown in FIG. 119(a), inwhich a sync signal S is transmitted by a time slot 780a andtransmission signals to respective terminal stations of A, B, C channelsby time slots 780b, 780c, 780d respectively at a frequency A. On theother hand, an up transmission, i.e. from the mobile station to the basestation, transmission is performed in such a manner that a sync signalS, and transmission signals of a, b, c channels are transmitted by timeslots 781a, 781b, 781c, 781d at a frequency B.

The present invention, which is characterized by a multi-level, e.g. 64SRQAM, signal transmission system, allows to have three-level dataconsisting of D₁, D₂, D₃ of 2 bit/Hz as shown in FIG. 119(b). As both ofA₁ and A₂ data are tranmitted by 16 SRQAM, their time slots have twotimes data rate as shown by slots 782b, 782c and 783b, 783c. It meansthe same quality sound can be transmitted by a half time. Accordingly, atime width of respective time slots 782b, 782c becomes a half. In thismanner, two times transmission capacity can be acquired at the two-levelregion 776c shown in FIG. 118, i.e. at the vicinity of the base station.

In the same way, time slots 782g, 783g carry out thetransmission/reception of E1 data by use of a 64 SRQAM signal. As thetransmission capacity is three times, one time slot can be used forthree channels of E₁, E₂, E₃. This would be used for an area furtherclose to the base station. Thus, up to three times communicationcapacity can be obtained at the same frequency band. An actualtransmission efficiency, however, would be reduced to 90%. It isdesirable for enhancing the effect of the present invention to coincidethe transmission amount distribution according to the present inventionwith the regional distribution of the actual traffic amount as perfectperfectly as possible.

In fact, an actual urban area consists of a crowded building districtand a greenbelt zone surrounding this building area. Even an actualsuburb area consists of a residential district and fields or a forestsurrounding this residential district. These urban and suburb areasresemble the distribution of the TF diagram. Thus, the application ofthe present invention will be effective.

FIG. 120 is a diagram showing time slots by the TDMA method, whereinFIG. 120(a) shows a conventional method and FIG. 120(b) shows thepresent invention. The conventional method uses time slots 786a, 786bfor transmission to portable phones of A, B channels at the samefrequency and time slots 787a, 787b for transmission from the same, asshown in FIG. 120(a).

On the contrary, 16 SRQAM mode of the present invention uses a time slot788a for reception of A₁ channel and a time slot 788c for transmissionto A₁ channel as shown in FIG. 120(b). A width of the time slot becomesapproximately ½. In case of 64 SRQAM mode, a time slot 788i is used forreception of D₁ channel and a time slot 788i is used for transmission toD₁ channel. A width of the time slot becomes approximately ⅓.

In order to save electric power, a transmission of E₁ channel isexecuted by use of a normal 4 SRQAM time slot 788r while reception of E₁channel is executed by use of a 16 SRQAM time slot 788p being a ½ timeslot. Transmission power is surely suppressed, although communicationcost may increase due to a long occupation time. This will be effectivefor a small and light portable telephone equipped with a small batteryor when the battery is almost worn out.

As is described in the foregoing description, the present inventionmakes it possible to determine the distribution of transmission capacityso as to coincide with an actual traffic distribution, therebyincreasing substantial transmission capacity. Furthermore, the presentinvention allows base stations or terminal stations to freely select oneamong two or three transmission capacities. If the frequency utilizationefficiency is lowered, power consumption will be decreased. If thefrequency utilization efficiency is selected higher, communication costwill be saved. Moreover, adoption of a 4 SRQAM having smaller capacitywill simplify the circuitry and reduce the size and cost of thetelephone. As explained in the previous embodiments, one characteristicsof the present invention is that compatibility is maintained among allof associated stations. In this manner, the present invention not onlyincreases transmission capacity but allows to provide customers a widevariety of series from a super mini telephone to a high performancetelephone.

Embodiment 9

Hereinafter, a ninth embodiment of the present invention will bedescribed referring to the drawings. The ninth embodiment employs thisinvention in an OFDM transmission system. FIG. 123 is a block diagram ofan OFDM transmitter/receiver, and FIG. 124 is a diagram showing aprinciple of an OFDM action. An OFDM is one of FDM and has a betterefficiency in frequency utilization as compared with a general FDM,because an OFDM sets adjacent two carriers to be quadrate with eachother. Furthermore, OFDM can bear multipath obstruction such as ghostand, therefore, may be applied in the future to the digital musicbroadcasting or digital TV broadcasting.

As shown in the principle diagram of FIG. 124, OFDM converts an inputsignal by a serial to parallel converter 791 into a data being disposedon a frequency axis 793 at intervals of 1/ts, so as to producesubchannels 794a^(˜) 794e. This signal is inversely FFT converted by amodulator 4 having an inverse FFT 40 into a signal on a time axis 799 toproduce a transmission signal 795. This inverse FFT signal istransmitted during an effective symbol period 796 of the time period ts.A guard interval 797 having an amount tg is provided between symbolperiods. -A transmitting/receiving action of HDTV signal in accordancewith this ninth embodiment will be explained referring to the blockdiagram of FIG. 123, which shows a hybrid OFDM-CCDM system. An inputtedHDTV signal is separated by a video encoder 401 into three-level levels,a low frequency band D₁₋₁, a medium-low frequency band D₁₋₂, and ahigh-medium-low frequency band D₂, video signals, and fed to an inputsection.

In a first data stream input 743, D₁₋₁ signal is ECC encoded with highcode gain and D₁₋₂ signal is ECC coded with normal code gain. A TDM 743743c performs time division multiplexing of D₁₋₁ and D₁₋₂ signals toproduce a D₁ signal, which is then fed to a D₁ serial to parallelconverter 791d in a modulator 852a. The D₁ signal consists of n piecesof parallel data, which are inputted into first inputs of n pieces ofC-CDM modulator 4a, 4b, - - - respectively.

On the other hand, the high frequency band signal D₂ is fed into asecond data stream input 744 of the input section 742, in which D₂signal is ECC (Error Correction Code) encoded in an ECC 744a and thenTrellis encoded in a Trellis encoder 744b. Thereafter, the D₂ signal issupplied to a D₂ serial to parallel converter 791b of the modulator 852aand converted into n pieces of parallel data, which are inputted intosecond inputs of the n pieces of C-CDM modulator 4a, 4b, - - -respectively.

The C-CDM modulators 4a, 4b, 4c - - - respectively produces produce 16SRQAM signal on the basis of D₁ data of the first data stream input andD₂ data of the second data stream input. These n pieces of C-CDMmodulator respectively has have a carrier different from each other. Asshown in FIG. 124, carriers 794a, 794b, 794c, - - - are arrayed on thefrequency axis 793 so that adjacent two carriers are 90°-out-of-phasewith each other. Thus C-CDM modulated n pieces of modulated signal arefed into the inverse FFT circuit 40 and mapped from the frequency axisdimension 793 to the time axis dimension 709 799. Thus, time signals796a, 796b - - - , having an effective symbol length ts, are produced.There is provided a guard interval zone 797a of Tg seconds between theeffective symbol time zones 796a and 796b, in order to reduce multipathobstruction. FIG. 129 is a graph showing a relationship between timeaxis and signal level. The guard time Tg of the guard interval band 797ais determined by taking account of multipath affection and usage ofsignal. By setting the guard time Tg longer than the multipath affectedtime, e.g. TV ghost, modulated signals from the inverse FFT circuit 40are converted by a parallel to serial converter 4e into one signal and,then, transmitted from a transmitting circuit 5 as an RF signal.

Next, an action of a receiver 43 will be described. A received signal,shown as time-base symbol signal 796e of FIG. 124, is fed into an inputsection 24 of FIG. 123. Then, the received signal is converted into adigital signal in a demodulator 852b and further changed into Fouriercoefficients in a an FFT 40a. Thus, the signal is mapped from the timeaxis 799 to the frequency axis 793a as shown in FIG. 124. That is, thetime-base symbol signal is converted into frequency-base carriers 794a,794b, - - - . As these carriers are in quadrature relationship with eachother, it is possible to separate respective modulated signals. FIG.125(b) shows thus demodulated 16 SRQAM signal, which is then fed torespective C-CDM demodulators 45a, 45b, - - - of a C-CDM demodulator 45,in which demodulated 16 SRQAM signal is demodulated into multi-level subsignals D₁, D₂. These sub signals D₁ and D₂ are further demodulated by aD₁ parallel to serial converter 852a and a D₂ parallel to serialconverter 852b into original D₁ and D₂ signals.

Since the signal transmission system is of C-CDM multi-level shown in125(b), both D₁ and D₂ signals will be demodulated under betterreceiving condition but only D₁ signal will be demodulated under worse,e.g. low C/N rate, receiving condition. Demodulated D₁ signal isdemodulated in an output section 757. As the D₁₋₁ signal has higher ECCcode gain as compared with the D₁₋₂ signal, an error signal of the D₁₋₁signal is reproduced even under worse receiving condition.

The D₁₋₁ signal is converted by a 1-1 video decoder 402c into a lowfrequency band signal and outputted as an LDTV, and the D₁₋₂ signal isconverted by a 1-2 video decoder 402d into a medium frequency bandsignal and outputted as EDTV.

The D₂ signal is Trellis decoded by a Trellis decoder 759b and convertedby a second video decoder 402b into a high frequency band signal andoutputted as an HDTV signal. Namely, an LDTV signal is outputted in caseof the low frequency band signal only. An EDTV signal of a wide NTSCgrade is outputted if the medium frequency band signal is added to thelow frequency band signal, and an HDTV signal is produced by adding low,medium, and high frequency band signals. As well as the previousembodiment, a TV signal having a picture quality depending on areceiving C/N rate can be received. Thus, the ninth embodiment realizesa novel multi-level signal transmission system by combining an OFDM anda C-CDM, which was not obtained by the OFDM alone.

An OFDM is certainly strong against multipath such as TV ghost becausethe guard time Tg can absorb an interference signal of multipath.Accordingly, the OFDM is applicable to the digital TV broadcasting forautomotive vehicle TV receivers. Meanwhile, no OFDM signal is receivedwhen the C/N rate is less than a predetermined value because its signaltransmission pattern is non not of a multi-level type.

However the present invention can solve this disadvantage by combiningthe OFDM with the C-CDM, thus realizing a gradational degradationdepending on the C/N rate in a video signal reception without beingdisturbed by multipath.

When a TV signal is received in a compartment of vehicle, not only thereception is disturbed by multipath but the C/N rate is deteriorated.Therefore, the broadcast service area of a TV broadcast station will notbe expanded as expected if the countermeasure is only for multipath.

On the other hand, a reception of TV signal of at least LDTV grade willbe ensured by the combination with the multi-level transmission C-CDMeven if the C/N rate is fairly deteriorated. As a picture plane size ofan automotive vehicle TV is normally less than 10 inches, a TV signal ofan LDTV grade will provide a satisfactory picture quality. Thus, theLDTV grade service area of automotive vehicle TV will be largelyexpanded. If an OFDM is used in an entire frequency band of HDTV signal,present semiconductor technologies cannot prevent circuitry scale fromincreasing so far.

Now, an OFDM method of transmitting only D₁₋₁ of low frequency band TVsignal will be explained below. As shown in a block diagram in FIG. 138,a medium frequency band component D₁₋₂ and a high frequency bandcomponent D₂ of an HDTV signal are multiplexed in C-CDM modulator 4a,and then transmitted at a frequency band A through an FDM 40d.

On the other hand, a signal received by a receiver 43 is first of allfrequency separated by an FDM 40e and, then, demodulated by a C-CDMdemodulator 4b of the present invention. Thereafter, thus C-CDMdemodulated signal is reproduced into medium and high frequencycomponents of HDTV in the same way as in FIG. 123. An operation of avideo decoder 402 is identical to that of embodiments 1, 2, and 3 andwill no more be not be further explained.

Meanwhile, the D₁₋₁ signal, a low frequency band signal of MPEG 1 gradeof HDTV, is converted by a serial to parallel converter 791 into aparallel signal and fed to an OFDM modulator 852c, which executes QPSKor 16 QAM modulation. Subsequently, the D₁₋₁ signal is converted by aninverse FFT 40 into a time-base signal and transmitted at a frequencyband B through a FDM 40d.

On the other hand, a signal received by the receiver 43 is frequencyseparated in the FDM 40e and, then, converted into a number offrequency-base signals in an FFT 40a of an OFDM modulator demodulator852d. Thereafter, frequency-base signals are demodulated in respectivedemodulators 4a, 4b, - - - and are fed into a parallel to serialconverter 882a, wherein a D₁D₁₋₁ signal is demodulated. Thus, a D₁₋₁signal of LDTV grad is outputted from the receiver 43.

In this manner, only an LDTV signal is OFDM modulated in the multi-levelsignal transmission. The method of FIG. 138 makes it possible to providea complicated OFDM circuit only for an LDTV signal. A bit rate of LDTVsignal is {fraction (1/20)} of that of an HDTV. Therefore, the circuitscale of the OFDM will be reduced to {fraction (1/20)}, which results inan outstanding reduction of overall circuit scale.

An OFDM signal transmission system is strong against multipath and willsoon be applied to a moving station, such as a portable TV, anautomotive vehicle TV, or a digital music broadcast receiver, which isexposed under strong and variable multipath obstruction. For such usagesa small picture size of less than 10 inches, 4 to 8 inches, is themainstream. It will be thus guessed that the OFDM modulation of a highresolution TV signal such as HDTV or EDTV will bring less effect. Inother words, the reception of a TV signal of LDTV grade would besufficient for an automotive vehicle TV.

On the contrary, multipath is constant at a fixed station such as a homeTV. Therefore, a countermeasure against multipath is relatively easy.Less effect will be brought to such a fixed station by OFDM unless it isin a ghost area. Using OFDM for medium and high frequency bandcomponents of HDTV is not advantageous in view of present circuit scaleof OFDM which is still large.

Accordingly, the method of the present invention, in which OFDM is usedonly for a low frequency band TV signal as shown in FIG. 138, can widelyreduce the circuit scale of the OFDM to less than {fraction (1/10)}without losing inherent OFDM effect capable of largely reducing multipleobstruction of LDTV when received at a mobile station such as anautomotive vehicle.

Although the OFDM modulation of FIG. 138 is performed only for D₁₋₁signal, it is also possible to modulate both D₁₋₁ and D₁₋₁D₁₋₂ by OFDM.In such a case, a C-CDM two-level signal transmission is used fortransmission of D₁₋₁ and D₁₋₂. Thus, a multi-level broadcasting beingstrong against multipath will be realized for a vehicle such as anautomotive vehicle. Even in a vehicle, the gradational graduation willbe realized in such a manner that LDTV and SDTV signals are receivedwith picture qualities depending on receiving signal level or antennasensitivity.

The multi-level signal transmission according to the present inventionis feasible in this manner and produces various effects as previouslydescribed. Furthermore, if the multi-level signal transmission of thepresent invention is incorporated with an OFDM, it will become possibleto provide a system strong against multipath and to alter datatransmission grade in accordance with receivable signal level change.

FIG. 126(a) shows another method of realizing the multi-level signaltransmission system, wherein the subchannels 794a-794c of the OFDM areassigned to a first layer 801a and the subchannels 794d-794f areassigned to a second layer 801b. There is provided a frequency guardzone 902a of f_(g) between these two, first and second, layers. FIG.126(b) shows an electric power difference 802b of Pg which is providedto differentiate the transmission power of the first and second layers801a and 801b.

Utilization of this differentiation makes it possible to increaseelectric power of the first layer 801a in the range not obstructing theanalogue TV broadcast service as shown in FIG. 108(d) previouslydescribed. In this case, a threshold value of the C/N ratio capable ofreceiving the first layer 801a becomes lower than that for the secondlayer 801b as shown in FIG. 108(e). Accordingly, the first layer 801acan be received even in a low signal-level area or in a large-noisearea. Thus, a two-layer signal transmission is realized as shown in FIG.147. This is referred to as Power-Weighted-OFDM system (i.e. PW-OFDM) inthis specification. If this PW-OFDM system is combined with the C-CDMsystem previously explained, three layers will be realized as shown inFIG. 108(e) and, accordingly, the signal receivable area will becorrespondingly expanded.

FIG. 144 shows a specific circuit, wherein the first layer data passingthrough the first data stream circuit 791a is modulated into thecarriers f₁-f₃ by the modulators 4a-4c having large amplitude and, then,are OFDM modulated in the inverse FFT 40. On the contrary, the secondlayer data passing through the second data stream circuit 791b ismodulated into the carriers f₆-f_(g) by the modulators 4d-4f havingordinary amplitude and, then, are OFDM modulated in the inverse FFT 40.Then, these OFDM modulated signals are transmitted from the transmitcircuit 5.

A signal received by the receiver 43 is separated into several signalshaving carriers of f₁-f_(n) through the FFT 40a. The carriers f₁-f₃ aredemodulated by the demodulators 45a-45c to reproduce the first datastream D₁, i.e. the first layer 801a. On the other hand, the carriersf₆-f₈ are demodulated by the demodulators 45d-45f to reproduce thesecond data stream D₂, i.e. the second layer 801b.

The first layer 801a has so large electric power that it can be receivedeven in a weak-signal area. In this manner, the PW-OFDM system realizesthe two-layer multi-level signal transmission. If this PW-OFDM iscombined with the C-CDM, it will become possible to provide 3-4 layers.As the circuit of FIG. 144 is identical with the circuit of FIG. 123 inthe remaining operations and, therefore, will no more not be furtherexplained.

Next, a method of realizing a multi-level signal transmission inTime-Weighted-OFDM (i.e. TW-OFDM) in accordance with the, presentinvention will be explained. Although the OFDM system is accompaniedwith the guard time zone t_(g) as previously described, adverseaffection of ghost will be eliminated if the delay time t_(M) of theghost, i.e. multipath, signal satisfies the requirement of t_(M)<t_(g).The delay time t_(M) will be relatively small, for example in the rangeof several μs, in a fixed station such as a TV receiver used for homeuse. Furthermore, as its value is constant, cancellation of ghost willbe relatively easily done. On the contrary, reflected wave will increasein case of a mobile station such as a vehicle TV receiver. Therefore,the delay time t_(M) becomes relatively large, for example in the rangeof several tens ps. Furthermore, the magnitude of t_(M) varies inresponse to the running movement of the vehicle. Thus, cancellation ofghost tends to be difficult. Hence, the multi-level signal transmissionis key or essential for such a mobile station TV receiver in order toeliminate adverse affection of multipath.

The multi-level signal transmission in accordance with the presentinvention will be explained below. A symbol contained in the subchannellayer A can be intensified against the ghost by setting a guard timet_(ga) of the layer A to be larger than a guard time t_(gb) of the layerB as shown in FIG. 146. In this manner, the multi-layer signaltransmission can be realized against multipath by use of weighting ofguard time. This system is referred to as Guard-Time-Weighted-OFDM (i.e.QTW-OFDM).

If the symbol number of the symbol time Ts is not different in the layerA and in the layer B, a symbol time t_(S3) of the layer A is set to belarger than a symbol time t_(sb) of the layer B. With thisdifferentiation, a carrier width Δfa of the carrier A becomes largersmaller than a carrier width Δfb of the carrier B. (Δfa> <Δfb)Therefore, the error rate becomes lower in the demodulation of thesymbol of the layer A compared with the demodulation of the symbol ofthe layer B. Thus, the differentiation of the layers A and B in theweighting of the symbol time Ts can realize a two-layer signaltransmission against multipath. This system is referred to asCarrier-Spacing-Weighted-OFDM (i.e. CSW-OFDM).

By realizing the two-layer signal transmission based on the GTW-OFDM,wherein a low-resolution TV signal is transmitted by the layer A and ahigh-frequency component is transmitted by the layer B, the vehicle TVreceiver can stably receive the low-resolution TV signal regardless oftough ghost. Furthermore, the multi-level signal transmission withrespect to the C/N ratio can be realized by differentiating the symboltime t_(s) based on the CSW-OFDM between the layers A and B. If thisCSW-OFDM is combined with the GTW-OFDM, the signal reception in thevehicle TV receiver can be further stabilized. High resolution is notnormally required to the vehicle TV or the portable TV.

As the time ratio of the symbol time including a low-resolution TVsignal is small, an overall transmission efficiency will not decrease somuch even if the guard time is enlarged. Accordingly, using the GTW-OFDMof the present invention for suppressing multipath by laying emphasis onthe low-resolution TV signal will realize the multi-layer type TVbroadcast service wherein the mobile station such as the portable orvehicle TV receiver can be compatible with the stationary station suchas the home TV without substantially lowering the transmissionefficiency. If combined with the CSW-OFDM or the C-CDM as describedpreviously, the multi-layer to the C/N ratio can be also realized. Thus,the signal reception in the mobile station will be further stabilized.

An affection of the multipath will be explained in more detail. In caseof multipaths 810a, 810b, 810c, and 810d having shorter delay time asshown in FIG. 145(a), the signals of both the first and second layerscan be received and therefore the HDTV signal can be demodulated. On thecontrary, in case of multipaths 811a, 811b, 811c, and 811d having longerdelay time as shown in FIG. 145(b), the B signal of the second layercannot be received since its guard time t_(gb) is not sufficiently long.However, the A signal of the first layer can be received without beingbothered by the multipath since its guard time t_(ga) is sufficientlylong. As described above, the B signal includes the high-frequencycomponent of TV signal. The A signal includes the low-frequencycomponent of TV signal. Accordingly, the vehicle TV can reproduce theLDTV signal. Furthermore, as the symbol time Tsa is set larger thansymbol time Tsb, the first layer is strong against deterioration of C/Nratio.

Such a discrimination of the guard time and the symbol time is effectiveto realize two-dimensional multi-layer signal transmission of the OFDMin a simple manner. If the discrimination of guard time is combined withthe C-CDM in the circuit shown in FIG. 123, the multi-layer signaltransmission effective against both multipath and deterioration of C/Nratio will be realized.

Next, a specific example will be described below.

The smaller the D/U ratio of the receiving signal becomes, the largerthe multipath delay time T_(M) becomes. Because, the reflectedwave-increases compared with the direct wave. For example, as shown inFIG. 148, if the D/U ratio is smaller than 30 dB, the delay time T_(M)exceeds 30 μs because of increase of the reflected wave. Therefore, ascan be understood from FIG. 148, it will become possible to receive thesignal even in the worst condition if the Tg is set to be larger than 50μs.

Accordingly, as shown in detail in FIGS. 149(a) and l49(b), three groupsof first 801a, second 801b, and third 801c layers are assigned in a 2 msperiod of 1 sec TV signal. The guard times 797a, 797b, and 797c, i.e.Tga, Tgb, and Tgc, of these three groups are weighted to be, forexample, 50 ps, 5 μs, and 1 μs, respectively, as shown in FIG. 149(c).Thus, three-layer signal transmission effective to the multipath will berealized as shown in FIG. 150, wherein three layers 801a, 801b, and 801care provided.

If the GTW-QFDM is applied to all the picture quality, it is doubtlessthat the transmission efficiency will decrease. However, if the GTW-OFDMis only applied to the LDTV signal including less information for thepurpose of suppression of multipath, it is expected that an overalltransmission efficiency will not be worsened so much. Especially, as thefirst layer 801a has a long guard time Tg of 50 μs, larger than 30 μs,it will be received even by the vehicle TV receiver. The circuit shownin FIG. 127 will be suitable for this purpose. Especially, therequirement to the quality of vehicle TV is LDTV grade. Therefore, itstransmission capacity will be approximately 1 Mbps of MPEG 1 class. Ifthe symbol time 796a, i.e. Tsa, is set to be 200 μs, with respect to the2 ms period as shown in FIG. 149, the transmission capacity becomes 2Mbps. Even if the symbol rate is lowered less than half, anapproximately 1 Mbps capacity can be kept. Therefore, it is possible toensure picture quality of LDTV grade. Although the transmissionefficiency is slightly decreased, the error rate can be effectivelylowered by the CSW-QFDM in accordance with the present invention. If theC-CDM of the present invention is combined with the GTW-OFDM,deterioration of the transmission efficiency will be able to beeffectively prevented. In FIG. 149, the symbol times 796a, 796b, and796c of the same symbol number are differentiated to be 200 μs, 150 μs,and 100 μs, respectively. Accordingly, the error rate becomes high inthe order of the first, second, and third layers so as to realize themulti-layer signal transmission.

At the same time, the multi-layer signal transmission effective to C/Nratio can be realized. By combining the CSW-OFDM and the CSW-OFDM, atwo-dimensional multi-layer signal transmission is realized with respectto the multipath and the C/N ratio as shown in FIG. 151. As describedpreviously, it is possible to combine the CSW-OFDM and the C-CDM of thepresent invention for preventing the overall transmission efficiencyfrom being lowered. In the first, 1-2, and 1-3 layers 801a, 851a, and851az, the LDTV grade signal can be stably received by, for example, thevehicle TV receiver subjected to the large multipath T_(M) and low C/Nratio. In the second and 2-3 layers 801b and 851b, thestandard-resolution SDTV grade signal can be received by the fixed orstationary station located, for example, in the fringe of the servicearea which is generally subjected to the lower C/N ratio and ghost. Inthe third layer 801c which occupies more than half of the service area,the HDTV grade signal can be received since the C/N ratio is high andthe ghost is less because of large direct wave. In this manner, atwo-dimensional multi-layer broadcast service effective to both the C/Nratio and the multipath can be realized by the combination of theGTW-OFDM and the C-CDM or the combination of the GTW-OFDM and theCSW-C-CDM in accordance with the present invention. Thus, the presentinvention realizes a two-dimensional, matrix type, multi-layer signaltransmission system effective to both the C/N ratio and thel themultipath, which has not ever been realized by the prior arttechnologies.

A timing chart of a three level (HDTV, SDTV, LDTV) television signal ina two-dimensional multilevel broadcast of three C/N levels and threemultipath levels is shown in FIG. 152. As shown in the figure, the LDTVsignal is positioned in slot 796a1 of the first level of level layer A,the level with the greatest resistance to multipath interference; theSDT synchronization signal, address signal, and other important highpriority signals are positioned in slot 796a2, which has the nextgreatest resistance to multipath interference, and slot 796b1, which hasstrong resistance to C/N deterioration. The MTV common signal, i.e., lowpriority signals, and HDTV high priority signals are positioned inlevels 2 and 3 of level B. SDTV, EDTV, HDTV, and other high frequencycomponent television signals are positioned in levels l, 2, and 3 oflevel C.

As the resistance to C/N deterioration and multipath interferenceincreases, the transmission raze rate drops, causing the TV signalresolution to drop, and achieving the three-dimensional gracefuldegradation effect shown in FIG. 153 and unobtainable with conventionalmethods. As shown in FIG. 153, the three-dimensional multilevelbroadcast structure of the invention is achieved with three parameters;C/N ratio, multipath delay time, and the transmission rate.

The present embodiment has been described using the example of atwo-dimensional multilevel broadcast structure obtained by combiningGTW-OFDM of the invention with C-CDM of the invention as previouslydescribed, or combining GTW-OFDM, CSW-C-CDM, but other two-dimensionalmultilevel broadcast structures can be obtained by combining GTW-OFDMand power-weighted OFDM, or GTW-OFDM with other C/N ratio multileveltransmission methods. FIG. 154 is obtained by transmitting the power ofcarriers 794a, 794c, and 794e with less weighting compared with carriers794b, 794d, and 794f, achieving a two level power-weighted OFDM. Twolevels are obtained by power weighting carriers 795a and 795c, which areperpendicular to carrier 794a, to carriers 795b and 795d. While a totalof four levels are obtained, the embodiment having only two levels isshown in FIG. 154. As shown in the figure, because the carrierfrequencies are distributed, interference with other analogtransmissions on the same frequency band is dispersed, and there isminimal adverse effect.

By using a time positioning varying the time width of guard times 797a,797b, and 797c for each symbol 796a, 796b, and 796c as shown in FIG.155, three-level multipath multilevel transmission can be achieved.Using the time positioning shown in FIG. 155, the A-, B-, and C-leveldata is distributed on the time axis. As a result, even if burst noiseproduced at a specific time occurs, data destruction can be preventedand the TV signal can be stably demodulated by interleaving the datafrom the different layers. In particular, by interleaving with the Alevel data distributed, interference from burst noise generated by theignition systems of other vehicles can be significantly reduced inmobile TV receivers.

Block diagrams of a specific ECC encoder 744j and a specific ECC decoder749j 759j are shown in FIG. 160a and FIG. 160b, respectively. FIG. 167is a block diagram of the deinterleaver 936b. The interleave table 954processed in the deinterleaver RAM 936a of the deinterleaver 936b isshown in FIG. 168a, and interleave distance L1 is shown in FIG. 168b.

Burst noise interference can be reduced by interleaving the data in thisway. By using a 4-level VSB, 8-level VSB, or 16-level VSB transmissionapparatus as described in embodiments 4, 5, and 6, respectively, andshown in the VSB receiver block diagram (FIG. 161) and the VSBtransmitter block diagram (FIG. 162), or by using a QAM or PSKtransmission apparatus as described in embodiments 1 and 2,respectively, burst noise interference can be reduced, and televisionreception with very low noise levels can be achieved in ground stationbroadcasting.

By using 3-level broadcasting by means Of of the method shown in FIG.155, LDTV grade television reception by mobile receivers, includingmobile TV receivers in motor vehicles and hand-held portable televisionsets, can be stabilized because level A has the effect of reducing burstnoise interference in addition to multipath interference and C/N ratiodeterioration.

The multi-level signal transmission method of the present invention isintended to increase the utilization of frequencies but may be suitedfor not all the transmission systems since causing some type receiversto be declined in the energy utilization, it is a good idea for use witha satellite communications system for selected subscribers to employmost advanced transmitters and receivers designed for best utilizationof applicable frequencies and energy. Such a specific purpose signaltransmission system will not be bound by the present invention.

The present invention will be advantageous for use with a satellite orterrestrial broadcast service which is essential to run in the samestandards for as long as 50 years. During the service period, thebroadcast standards must not be altered but improvements will beprovided time to time corresponding to up-to-date technologicalachievements. Particularly, the energy for signal transmission willsurely be increased on any satellite. Each TV station should provide acompatible service for guaranteeing TV program signal reception to anytype receivers ranging from today's common ones to future advanced ones.The signal transmission system of the present invention can provide acompatible broadcast service of both the existing NTSC and HDTV systemsand also, ensure a future extension to match mass date datatransmission.

The present invention concerns much on the frequency utilization thanthe energy utilization. The signal receiving sensitivity of eachreceiver is arranged different differently depending on a signal statelevel to be received so that the transmitting power of a transmitterneeds not be increased largely. Hence, existing satellites which offer asmall energy for reception and transmission of a signal can best be usedwith the system of the present invention. The system is also arrangedfor performing the same standards corresponding to an increase in thetransmission energy in the future and offering the compatibility betweenold and new type receivers. In addition, the present invention will bemore advantageous for use with the satellite broadcast standards.

The multi-level signal transmission method of the present invention ismore preferably employed for terrestrial TV broadcast service in whichthe energy utilization is not crucial, as compared with satellitebroadcast service. The results are such that the signal attenuatingregions in a service area which are attributed to a conventional digitalHDTV broadcast system are considerably reduced in extension and also,the compatibility of an HDTV receiver or display with the existing NTSCsystem is obtained. Furthermore, the service area is substantiallyincreased so that program suppliers and sponsors can appreciate moreviewer viewers. Although the embodiments of the present invention referto 16 and 32 QAM procedures, other modulation techniques including 64,128, and 256 QAM will be employed with equal success. Also, multiplePSK, ASK, and FSK techniques will be applicable as described with theembodiments.

A combination of the TDM with the SRQAM of the present invention hasbeen described in the above. However, the SRQAM of the present inventioncan be combined also with any of the FDM, CDMA and frequency dispersalcommunications systems.

What is claimed is:
 1. A digital TV receiver comprising: a receivingsection for receiving a PSK (Phase Shift Key) modulation signalcomprising a plurality of signal points disposed on specific phases of agiven constellation in a signal space diagram; a demodulator fordemodulating the received signal from said receiving section into adigital, signal; an error correcting section for error correcting ademodulation signal from said demodulator; and an image expander farexpanding an error-corrected signal from said error correcting sectionto a video signal, thereby outputting a video signal, wherein saiddemodulator demodulates the received signal as first and second PSKsignals, said first PSK signal representing a first data stream to bereproduced, said first PSK signal comprising n-value signal points, saidsecond PSK signal representing both said first data stream and a seconddata steam to be reproduced, said second PSK signal comprising m-valuesignal points, where m is an integer larger than n, wherein the valuesignal points of said second PSK signal are divisible into a groups ofsignal points which are distinguishable from one another in the signalspace, and said demodulator distinguishes said n groups of signal pointsfrom one another as the n-value signal points of sold first PSK signalby demodulating the received signal as said first PSK signal, andwherein high priority information is demodulated at least in said firstdata stream.
 2. The digital TV receiver in accordance with claim 1,wherein said demodulator demodulates information relating to alow-resolution component of the video signal from said first data streamand demodulates information relating to a high-resolution component ofthe video signal from said second data stream.
 3. The digital TVreceiver in accordance with claim 1, comprising means for stoppingoutput of said second data stream when an error rate of the receivedsignal is high.
 4. The digital TV receiver in accordance with claim 1,wherein said demodulator comprises first means for demodulating thereceived signal as a QPSK signal to reproduce the first data stream andsecond means for demodulating the received signal as a SPSK signal toreproduce both of the first data stream and the second data stream.
 5. Adigital TV receiver comprising: a receiving section for receiving a PSK(Phase Shift Key) modulation signal comprising m-value signal pointsdisposed on specific phases of a given constellation in a signal spacediagram and representing a first data stream and a second data stream tobe reproduced; a demodulator for demodulating said PSK modulation signalfrom said receiving section into a digital signal; an error correctingsection for error correcting a demodulation signal from saiddemodulator; and an image expander for expanding an error-correctedsignal from said error correcting section to a video signal, whereinsaid demodulator includes means for demodulating said PSK signalcomprising m-value signal points as another PSK signal, said another PSKsignal comprising n-value signal points and representing only said firstdata stream, where n is an integer smaller than m, and wherein highpriority information is demodulated at least in said first: data stream.6. A signal transmission system comprising a transmission apparatus anda receiving apparatus, said transmission apparatus comprising: a trellisencoder operable to encode a source signal to produce an encoded signal;a modulator operable to modulate the encoded signal to produce an8-level VSB modulated signal; and a transmitter operable to transmit the8-level VSB modulated signal in a 6 MHz bandwidth; said receivingapparatus comprising: a receiver operable to receive the transmitted8-level VSB modulated signal; a demodulator operable to demodulate the8-level VSB modulated signal, received by said receiver, to produce ademodulated signal; and a trellis decoder operable to decode thedemodulated signal to produce the source signal, wherein a code rate ofsaid trellis encoder is {fraction (2/3)} and a code rate of said trellisdecoder is {fraction (2/3)}.
 7. A signal transmission apparatuscomprising: a trellis encoder operable to encode a source signal toproduce an encoded signal; a modulator operable to modulate the encodedsignal to produce an 8-level VSB modulated signal; and a transmitteroperable to transmit the 8-level VSB modulated signal in a 6 MHzbandwidth, wherein a code rate of said trellis encoder is {fraction(2/3)}.
 8. A signal receiving apparatus comprising: a receiver operableto receive an 8-level VSB modulated signal, having information of asource signal, transmitted in a 6 MHz bandwidth; a demodulator operableto demodulate the 8-level VSB modulated signal, received by saidreceiver, to produce a demodulated signal; and a trellis decoderoperable to decode the demodulated signal to produce the source signal,wherein a code rate of said trellis decoder is {fraction (2/3)}.
 9. Asignal transmission and receiving method comprising: trellis encoding asource signal to produce an encoded signal; modulating the encodedsignal to produce an 8-level VSB modulated signal; transmitting, the8-level VSB modulated signal in a 6 MHz bandwidth; receiving thetransmitted 8-level VSB modulated signal; demodulating the 8-level VSBmodulated signal, received in said receiving, to produce a demodulatedsignal; and trellis decoding the demodulated signal to produce thesource signal, wherein a code rate of said trellis encoding is {fraction(2/3)} and a code rate of said trellis decoding is {fraction (2/3)}. 10.A signal transmission method comprising: trellis encoding, a sourcesignal to produce an encoded signal; modulating the encoded signal toproduce an 8-level VSB modulated signal; and transmitting the 8-levelVSB modulated signal in a 6 MHz bandwidth, wherein a code rate of saidtrellis encoding is {fraction (2/3)}.
 11. A signal receiving methodcomprising: receiving an 8-level VSB modulated signal, havinginformation of a source signal, transmitted in a 6 MHz bandwidth;demodulating the 8-level VSB modulated signal, received in saidreceiving, to produce a demodulated signal; and trellis decoding thedemodulated signal to produce the source signal, wherein a code rate ofsaid trellis decoding is {fraction (2/3)}.